Bidirectional bipolar-mode JFET driver circuitry

ABSTRACT

Double sided versions of several power transistor types are devices that are already known in the literature. Devices built in this configuration are generally required to have a separate driver circuit to control the front and rear control electrodes and provide the gate or base voltage and/or currents for the power switch. This is because there may be of the order of 1000V potential-difference between the frontside and rearside potentials when the transistor is in the off condition—and a single integrated circuit cannot generally sustain this within a single package. The NPN configuration is preferred in this case to benefit from electron conduction for the main power path between the emitters. However, problems arising when using a P-type wafer. The present invention seeks to avoid the use of P-type wafers while still getting the higher conduction performance of NPN operation.

CROSS-REFERENCE OF THE RELATED APPLICATIONS

This application is a continuation-in-part of U.S. patent applicationSer. No. 15/595,361, filed May 15, 2017, which claims priority to UnitedKingdom patent application number GB1621043.7 filed on Dec. 12, 2016 andis a continuation-in-part of U.S. patent application Ser. No.14/797,498, filed Jul. 13, 2015, which is a continuation-in-part ofInternational Application No. PCT/GB2014/050368 filed Feb. 7, 2014 whichclaims priority to United Kingdom patent application numbers GB1302196.9filed Feb. 7, 2013, GB1305250.1 filed on Mar. 22, 2013, GB1309465.1filed on May 27, 2013, GB1311298.2 filed on Jun. 25, 2013, GB1321151.1filed on Nov. 29, 2013, GB1322177.5 filed on Dec. 16, 2013, GB1400866.8filed on Jan. 20, 2014 and GB1412513.2 filed on Jul. 14, 2014, theentire disclosures of which are incorporated herein by reference intheir entirety; this application also claims priority to United Kingdompatent application number GB1807139.9 filed May 1, 2018. Alsoincorporated herein by reference in their entirety are European patentapplication number EP85106861.9 filed Jun. 4, 1985, United Kingdompatent application numbers GB1621043.7 filed Dec. 12, 2016 andGB1711544.5 filed Jul. 18, 2017, International Application No.PCT/GB2017/053713 filed Dec. 12, 2017 which claims priority to UnitedKingdom patent application numbers GB1621043.7, U.S. patent applicationSer. No. 14/514,878, filed Oct. 15, 2014, K. D. Hobart “Fabrication of adouble-side IGBT by very low temperature wafer bonding”, PowerSemiconductor Devices and ICs, 1999. ISPSD '99. Proceedings,Experimental Demonstration of High-Voltage 4H—SiC Bi-Directional IGBTs,S. Chowdhury, IEEE Electron Device Letters (Volume: 37, Issue: 8, August2016) pp 1033-1036, The bipolar mode FET: a new power device combiningFET with BJT operation, Microelectronics Journal Volume 24, Issues 1-2,January 1993, Pages 61-74, and De Doncker, R. W.; Lyons, J. P.; “Theauxiliary resonant commutated pole converter,” Industry ApplicationsSociety Annual Meeting, 1990, Conference Record of the 1990 IEEE, vol.,no., pp. 1228-1235 vol. 2, 7-12 Oct. 1990.

FIELD OF THE INVENTION

This invention relates to driving bidirectional semiconductor powertransistor devices, particularly but not exclusively, to driving adouble-sided, bidirectional Bipolar-Mode JFET (BMJFET).

BACKGROUND OF THE INVENTION

Double sided versions of several power transistor types are devices thatare already known in the literature e.g. bidirectional double sidedSilicon Carbide IGBTs as illustrated in FIG. 43, Thyristors and Triacs.Almost any vertical-power high voltage devices are capable ofbidirectionality by forming control features of such a device on boththe front and back surfaces and sharing a common drift region of lowlydoped silicon between the two sides of silicon.

Devices built in this configuration are generally required to have aseparate driver circuit to control the front and rear control electrodesand provide the gate or base voltage and/or currents for the powerswitch. This is because there may be of the order of 1000Vpotential-difference between the frontside and rearside potentials whenthe transistor is in the off condition—and a single integrated circuitcannot generally sustain this within a single package.

One double-side bidirectional device is described in granted U.S. Pat.No. 9,054,706 (referred to as the ‘BTRAN’ herein and reproduced in FIG.42 for convenience) and in its preferred embodiment is constructed in anNPN format using a P-type wafer with N+ emitter diffusions front andback sides. The NPN configuration is preferred in this case to benefitfrom electron conduction for the main power path between the emitterswhich is 2× higher conductivity than hole conduction as would be presentfor a PNP version

However, problems arising when using a P-type wafer include 1) Lack ofsuitable high-voltage P-type wafer: Only an N-type wafer can benefitfrom the NTD (Neutron Transmutation Doping) system that converts siliconinto phosphorus at a uniform concentration giving excellent uniformitywhich supports >3 kV operation. 2) While in theory, there should be noproblem producing a good quality P-type high voltage float-zone waferfor <3 kV operation, in practice these are not readily available becauseno market currently exists for them. 3) P-type wafers require differentfield termination structures and/or passivation requiring somedevelopment effort. 4) NPN devices constructed on P-wafers have anenhanced avalanche breakdown multiplication coefficient potentiallyreducing breakdown voltage. 5) For Silicon-Carbide devices, there is noknown method of controlling the P-type dopants to the accuracy requiredfor high voltage P-type wafers.

Alternatively, a PNP version of the BTRAN can be envisaged using anN-type wafer where the structure remains the same but the doping typesare swapped (P→N and N→P). Unfortunately a PNP type power structure usesholes as its main carries for current and therefore has a 2:1 reductionin conductivity in the saturation resistance region vs NPN. Also, in thecontext of Silicon Carbide, the even more severe reduction of holemobility in this material makes a PNP a poor choice.

SUMMARY OF THE INVENTION

The present invention seeks to avoid the use of P-type wafers whilestill getting the higher conduction performance of NPN operation.

According to a first aspect of the present invention, there is provideda method of driving a bidirectional semiconductor power transistordevice, the method comprising: driving the semiconductor powertransistor device on a high side of the transistor that is at a higherpotential than an opposing low side of the transistor, the high side ofthe transistor comprising a P+ electrode and a N+ electrode; andselectively steering current into either the P+ electrode of thesemiconductor power transistor device or into the N+ electrode of thesemiconductor power transistor device on the high side of thesemiconductor power transistor device.

The current being switched is that of the through current of the powerswitch, not an external power source, which isn't required.

In this way, their respective good turn on and good turn off lossproperties may be exploited.

Sequencing various switches may enable a bipolar ‘base’ drive current tonot require an external source of power and/or the device may be swapped(‘rolled’) between NPN and PNP drive.

Protection and control isolation features may be facilitated.

The method may further comprise using a controller to determine, througha voltage comparator, the side of the transistor that is at the lowestpotential and driving the semiconductor power transistor deviceaccordingly.

Wherein driving the semiconductor power transistor device may comprisedriving a gate of the semiconductor power transistor device.

Wherein the controller may drive the semiconductor power transistordevice by outputting a PWM signal to drive switches.

Wherein the switches may comprise low on-resistance mosfets.

Wherein the bidirectional semiconductor power transistor device maycomprise a double sided bidirectional semiconductor power transistordevice.

Wherein the bidirectional semiconductor power transistor device maycomprise a bidirectional Bipolar-Mode JFET (BMJFET).

Wherein the bidirectional Bipolar-Mode JFET (BMJFET) may comprise adouble sided bidirectional Bipolar-Mode JFET (BMJFET).

Wherein the bidirectional Bipolar-Mode JFET (BMJFET) may comprise anultra-lateral single sided bidirectional Bipolar-Mode JFET (BMJFET).

Wherein the bidirectional semiconductor power transistor device maycomprise a Silicon Carbide semiconductor construction.

Wherein the bidirectional semiconductor power transistor device maycomprise N-type wafer.

According to a second aspect of the present invention, there is providedan electronic component comprising: a bidirectional semiconductor powertransistor device; and a switching construction configured to drive thebidirectional semiconductor power transistor device in the mannerrecited in the first aspect.

Wherein the electronic component comprises an auxiliary resonantcommutated pole converter auxiliary switch.

Traditional power switches are fabricated from P and N type regions ofsemiconductor material with metalised layers and sometimes with thinoxide layers for field-effect devices. Well known in the art aretechnologies such as Bipolar Junction Transistors (BJT), Junction FieldEffect Transistors (JFET), Metal Oxide Field Effect Transistors (MOSFET)and Insulated Gate Biploar Transistor (IGBT) which generally switch DCvoltages. Whereas Triacs, Silicon Controlled Rectifiers (SCR) as well asmodified (typically back-to-back) versions of the DC switch types havebeen used to switch AC power. Whether fabricated on Silicon or acompound semiconductor such has Silicon Carbide (SiC) all these standarddevices has one or more limitation when it comes to the common task ofswitching AC power on and off

These limitations lead to the following problems.

Lack of AC switching ability.

Expensive semiconductor materials.

Complex semiconductor processing steps.

Inherent voltage drop on turn on (typ. 0.8 to 2.5 volts) limitingefficiency.

Lack of turn-off ability—can prevent implementation of short-circuitprotection feature.

Complex, high-voltage, high current +ve and −ve base/gate drivecircuitry.

High on-resistance for high voltage devices.

It is an object of the present invention to address the problemsdescribed above.

The proposed new device can be manufactured at low cost on standard BJTfabrication equipment, has inherent AC switching ability and simpledrive requirements. Voltage drop at turn on can be under 100 mV at 10'sof amps per cm·squared and the device can switch high voltages both onand off at will with a single polarity drive pulse without latch-up. Inconjunction with a low cost microcontroller, these switches can performefficient mains voltage switching, short circuit protection, loaddiagnostics and data logging functions for smart power, smart appliancesystems without requiring special heatsinks or fan cooling.

A bi-directional bipolar junction transistor (BJT) structure maycomprise: a base region of a first conductivity type, wherein said baseregion constitutes a drift region of said structure; first and secondcollector/emitter (CE) regions, each of a second conductivity typeadjacent opposite ends of said base region; wherein said base region islightly doped relative to said collector/emitter regions; the structurefurther comprising: base connection to said base region, wherein saidbase connection is within or adjacent to said first collector/emitterregion.

It will be appreciated that the term “drift region” refers to a highvoltage sustaining region including a relatively low dopingconcentration. During the off-state of the device, the drift region issubstantially fully depleted. In a conventional high voltage BJT, thecollector region generally acts as the high voltage sustaining layer. Onthe contrary, the thick base, drift region works as the high voltagesustaining layer. Furthermore, in the conventional high voltageThyristor, the main current conduction generally takes place throughfour active semiconductor layers, e.g. cathode (N), gate (P), ndrift (N)and anode (P). By contrast, the bi directional device of the presentinvention provides current conduction through three active layers, i.e.first CE region, base region and the second CE region.

Broadly speaking the base connection and base region define a secondtransistor which enables the bi-directional BJT structure to support ahigh voltage in a case where, in effect, a base emitter junction of thestructure is forward biased so that a high current through this junctionwould otherwise flow. This second transistor may either constitute afurther BJT or may be a JFET, depending upon whether the base connectionis within or in the vicinity of/adjacent to the first CE region,respectively. Thus where the base connection is within the first CEregion the second transistor is a BJT transistor with collector/emitterterminals connected to the base connection and to the base, drift regionrespectively and with its base connection formed by the first CE regionof the bi-directional BJT structure. Typically the first and second CEregions of the bi-directional BJT structure are heavily doped, forexample in a

range between 10.sup.18 cm·sup.−3 to 10.sup.21 cm·sup.−3 whilst the baseregion is lightly doped compared to the CE regions. The first CE regionmay extend laterally to form the base of the second transistor. Thus,for example, the first and second CE regions of the BJT structure may beN.sup.+ type and the base region of the bi-directional BJT structureP.sup.−, and the second transistor may then be a PNP transistor with thebase region comprising a preferably relatively lower doped, N.sup.−extension of the first, N.sup.+CE region, and with collector/emitterterminals of the second transistor comprising the P.sup.− base regionand the base connection. In preferred embodiments the base connection isan ohmic connection comprising a region of the first conductivity type,typically heavily doped, for example a range between 10.sup.18 cm·sup.−3to 10.sup.21 cm·sup.−, such as a P.sup.+ region.

In other embodiments the second transistor is a JFET and the baseconnection/ohmic contact (P+) and base, drift region constitutesource/drain connections of the JFET. The gate terminal of the JFET isthen formed by the (heavily doped, N.sup.+) first CE region of the bidirectional BJT structure. In this case the (ohmic) base connection isadjacent (which here includes slightly spaced apart from) the first CEregion. The channel region of the JFET lies between the (ohmic) baseconnection and the base drift region so that, in effect, the JFETcontrols conduction between the base connection and the base driftregion.

It can therefore be appreciated that where there is a forward conductionpath from the base region to the second CE region the second transistoreffectively stops substantially the entire voltage of what may be a highvoltage across the device appearing between the base connection and thesecond CE region—this would otherwise drive a large current through thedevice which would destroy it. Instead when a forward conduction pathbetween the base region and the second CE region is present (driven by avoltage on the base region relative to the second CE region) a forwardconduction path between the base connection and the base region includesa depleted portion of the base region—that is the base connection of thestructure is effectively isolated by the depleted portion of the baseregion. Thus during switch on, when a high voltage can appear across thefirst and second CE regions, substantially all this voltage can alsoappear between the base connection and second CE region across the basedepletion region. As the device turns on the voltage across the firstand second CE terminals will fall to a very low voltage and this ceasesto be a potential problem.

Preferably when the base region and the base connection are of p-typeand the first and second CE regions are of n-type, the following on andoff states occur in the BJT structure: (1) when no voltage is applied toany terminals, the structure may be in an off-state so as to formdepletion regions between said first CE region and base region andbetween said second CE

region and base region; (2) when a positive voltage is applied to saidsecond CE region and no voltage is applied to the first CE region andthe base connection, the structure is in an off-state so as to form adepletion region between said second CE region and base region; (3) whena negative voltage is applied to said second CE region and no voltage isapplied to the first CE region and the base connection, the structuremay be in an off-state so as to form a depletion region between saidfirst CE region and base region; (4) when a first positive voltage isapplied to said second CE region, a second positive voltage beingapplied to the base connection and no voltage is applied to the first CEregion, the structure is in an on-state in which majority carriers fromthe first CE region flow through the base region towards the second CEregion, and minority carriers from the base connection are injected intothe base region, the minority carriers being recombined with themajority carriers in a region adjacent the first CE region;

(5) when a negative voltage is applied to said second CE region, apositive voltage being applied to the base connection and no voltage isapplied to the first CE region, the structure is in an on-state in whichmajority carriers from the second CE region flow through the base regiontowards the first CE region, and minority carriers from the baseconnection are injected into the base region flowing towards the secondCE region, the minority carriers being recombined with the majoritycarriers in a region adjacent the second CE region. Here the majoritycarriers are electrons and the minority carriers are holes.

Preferably when the base region and the base connection are of n-typeand the first and second CE regions are of p-type, the following on andoff states occur in the BJT structure: (1) when no voltage is applied toany terminals, the structure may be in an off-state so as to formdepletion regions between said first CE region and base region andbetween said second CE region and base region; (2) when a positivevoltage is applied to said second CE region and no voltage is applied tothe first CE region and the base connection, the structure is in anoff-state so as to form a depletion region between said first CE regionand base region; (3) when a negative voltage is applied to said secondCE region and no voltage is applied to the first CE region and the baseconnection, the structure may be in an off-state so as to form adepletion region between said second CE region and base region; (4) whena first positive voltage is applied to said second CE region, a negativevoltage being applied to the base connection and no voltage is appliedto the first CE region, the structure is in an on-state in which holesfrom the second CE region flow through the base region towards thesecond CE region, and electron from the base connection are injectedinto the base region, the electrons being recombined with the holes in aregion adjacent the second CE region; (5) when a negative voltage isapplied to said second CE region, a negative voltage being applied tothe base connection and no voltage

is applied to the first CE region, the structure is in an on-state inwhich holes from the first CE region flow through the base regiontowards the second CE region, and electrons from the base connection areinjected into the base region flowing towards the first CE region, theelectrons being recombined with the holes in a region adjacent the firstCE region.

Conveniently, in some preferred embodiments the base connection isrecessed into a surface of the structure—conveniently an ohmicconnection of the first conductivity type can be formed within arecessed portion of the first CE region. Conveniently the recess may besufficient to also incorporate a metal connection to the ohmic forexample P.sup.+ region. It will be appreciated that in the proposeddevice the base region may be ohmic in nature so as to drive the maintransistor (not the second transistor) comprising the first CE region,base region and second CE region into the saturation region in whichboth saturation diffusion and drift current apply.

It will be appreciated that the labelling of the first and second CEregions is arbitrary. Although In some preferred embodiments the deviceis a vertical device, a lateral device may also be fabricated. Forexample the first and second CE regions may be fabricated in a commonlayer, displaced laterally from one another, separated by a base regionwhich runs beneath the first and second CE regions and laterally,joining these regions to one another.

In some preferred embodiments the base, drift region is wider in adirection between the ends of the region adjacent the CE regions thaneach CE region. However in some very high voltage devices the CE regionsthemselves may be relatively wide, comprising a long (deep) diffusion.In embodiments a current carrying capability of a connection pathbetween the base connection and the second CE region is less than acurrent carrying capability of a connection path between the first andsecond CE regions—that is the main conduction path through the device isbetween the first and second CE regions, the base connection merelyproviding a relatively small base current. It will further beappreciated that the structure is non-latching, that is a connectionbetween the first and second CE regions is switched off on removal of avoltage from the base connection (without requiring a current betweenthe first and second CE regions to go to zero).

A bipolar junction transistor (BJT) structure may comprise: a baseregion of a first conductivity type, wherein said base regionconstitutes a drift region of said structure; first and secondcollector/emitter (CE) regions, each of a second conductivity typeadjacent opposite ends of said base region; wherein said base region islightly doped relative to said collector/emitter regions; the structurefurther comprising: a base connection to said base region, wherein said

base connection is within or adjacent to said first collector/emitterregion and a buried layer of the second conductivity type disposedbetween the second CE region and the base region.

This bipolar junction transistor structure could be termed as anon-insulated gated bipolar junction transistor (NIGBT). It will beappreciated that the NIGBT devices preferably operate in DC applicationsin which the buried layer can help to sustain high voltage due to thepunch-through structure. The NIGBT structure is capable of operating inAC mode as well. However, in the AC mode, voltage sustaining capabilitycan be limited in one direction (or during a DC application) as thedevice try to deplete the highly doped buried layer.

A bipolar junction transistor (BJT) structure may comprise: a baseregion of a first conductivity type, wherein said base regionconstitutes a drift region of said structure, the drift region being areverse voltage sustaining region; an emitter region of a secondconductivity type; a collector of a second conductivity type, thecollector and emitter being adjacent opposite ends of said base region;wherein said base region is lightly doped relative to said collector andemitter regions; the structure further comprising: a base connectionregion of the first conductivity type formed adjacent to said emitterregion and a field stop layer of the first conductivity type formedbetween the emitter region and the base region, the base connectionbeing within the field stop layer.

The doping concentration of the field stop layer may be less than thatof the base connection. The thickness of the field stop layer may bemore than that of the base connection. The BJT structure may beconfigured such that a diode is formed between the collector and baseregion. The diode may be configured to operate as a reverse conductingdiode when driven by a driver circuit. The driver circuit may include asoftware controlled driver which can allow the BJT structure to provethe free-wheeling diode characteristics. The software controlled drivermay be configured such that the free-wheeling virtual diode may notallow large reverse voltage to occur.

The invention may provide a driver circuit operatively connected to theBJT structures above, the driver circuit comprising a first PWMcontroller and a second PWM controller, the first and second PWMcontrollers being coupled to one another, wherein the first PWMcontroller is capable of controlling a low voltage transistor and thesecond PWM controller is a high-frequency converter. The invention mayalso provide a driver circuit operatively connected to a standard BJTstructure—thus it would be apparent that the driver circuit described inthis specification are not limited to the proposed BJT structuresdisclosed but they are applicable to standard BJT structures as well.

The second PWM controller may be a buck converter using an inductor. Thecross modulation of the first and second PWM controllers across a basedrive inductance may allow a continuous dynamic current control of theBJT structure with high speed on and off capability. The frequency ofthe second PWM controller may be such that the off time is ofsubstantially similar order compared to the minority carrier (e.g.electron) lifetime ensuring that conductivity of the BJT structureremains substantially constant during the period controlled by thesecond PWM controller. The second PWM controller may be a phase offsetcontroller. The driver circuit may further comprise a third PWMcontroller which is a phase offset controller, wherein the second andthird phase offset controllers each drive an inductor with a commonpoint on a base connection terminal of a transistor. The first PWMcontroller may be coupled to the base connection terminal of thetransistor so as to drive the transistor. The multiphase offsets of thesecond and third PWM controllers create a relatively high effectivefrequency to match a reduced minority-carrier lifetime of high speedtransistors. The combination of the driver circuit and the BJT structuremay be configured to provide a reverse conducting diode.

When the base region is of p conductivity type, the driver may beconfigured to detect a negative current of the base so that holes can bepulled out from the junction between the first CE region and base toclamp a negative excursion.

The current rating of the base region substantially equal to that of thecollector or emitter so as to yield a reverse free-wheeling diode ofsubstantially equal current rating to the forward rating.

The base region may be configured to be clamped to ground using a MOSFETfrom the driver. The base current may be produced by the driver circuitto result in a reverse bias switch on transistor action so as to reducethe voltage drop in a current flow path below a normal voltage drop of adiode.

A computer program product comprising a computer readable medium inwhich a computer program is stored, the computer program comprisingcomputer readable code which, when run by a controller of a drivercircuit, causes the driver circuit to operate as the driver circuitabove, wherein the computer program is stored on the computer readablemedium.

When the controller is configured to detect a voltage of the BJTstructure in an off-state, the computer program product may configuredto turn on the BJT structure to emulate a reverse conducting diodeaction.

In a related aspect the invention provides a driver circuit, inparticular for a structure as described above, comprising a voltagesensing resistor and a current sensing resistor each coupled to one ofthe CE regions (optionally the same region). A microcontroller iscoupled to

the resistors to receive respective voltage sensing and current sensingsignals. In embodiments the microcontroller is configured to provide aPWM (pulse width modulation) output for controlling a current into thebase connection of the structure, via an inductance. This arrangementfacilitates accurate control of the device along a defined operatingpath. The voltage across the device and current through the device aresensed and the current into the base connection may then be adjusted tomove the device between a switched-on and a switched-off configurationin a controlled manner.

In a further aspect the invention also provides a circuit breakercomprising a first, power semiconductor switching device and a drivercircuit, wherein said circuit breaker has two power switching terminals,and further comprises a power supply, and a controller for said powersemiconductor switching device powered by said power supply, whereinsaid power supply is coupled in series with said first powersemiconductor switching device between said power switching terminals toderive a power supply from said terminals whilst said powersemiconductor switching device is on, and wherein said power supplycomprises a second switching device coupled in series with said firstpower semiconductor switching device, between said power switchingterminals, such that the circuit breaker is operable without a separatepower supply.

In embodiments, by employing a second switching device in series withthe first the relatively high current through the power switching devicewhen on can be leveraged with only a very small voltage drop to generatesufficient power for driving a base current into the device. It will beappreciated that when the power switching device is on the voltage dropacross the two power switching terminals should be as low aspracticable, and by employing a second, low voltage switching device inseries with the first a reasonable power, for example of order 1 wattcan be achieved with a very small voltage drop, for example of order 0.1volts, without wastage in, for example, a lead resistor. In preferredembodiments the power switching device is a high voltage device and thesecond device is a low voltage device, in particular forming part of theinput stage of a DC-to-DC converter. In this context a power devicerefers to a high voltage device which typically operates with a voltagein the range greater than 100 volts, 500 volts or 1000 volts and or atcurrents at greater than 1 amp, 10 amps or 100 amps. A low voltagedevice typically operates at a voltage of less than 50 volts, inembodiments less than 10 volts.

An aspect of the present invention provides, a method of etching asubstrate, wherein the substrate is a silicon substrate or a substratehaving a silicon surface, is disclosed. The method includes placing thesubstrate in a container, wherein the substrate is a N-type substrate;providing a volume of an acid solution in the container, wherein theacid solution serves as an

insulator; and drilling, using one or more needles supplied with avoltage, one or more holes on the surface of the substrate to locallyinvert the N-type substrate to a P-type substrate, wherein the voltageapplied on the surface of the substrate anodically etches the surface ofthe substrate to create the one or more holes by surface inversion.

It is preferable that the circuit breaker can be installed either wayaround in a circuit and thus the direction of current flow between thepower switching terminals may not be known. In preferred embodiments,therefore, the power supply is a switched mode power convertercomprising a plurality of low voltage switching devices arranged tocharge and discharge an energy storage component, (capacitor and/orinductor) so that power for the controller is provided with the sameplurality no matter which way round the circuit breaker is connectedinto each circuit. Thus the power supply may further comprise a sensorto sense a direction of current flow for controlling the plurality ofswitching devices according. Broadly speaking the switches are arrangedso that whichever the direction the current flows a positive side of theenergy storage component delivers power to a positive line for thecontroller, and vice versa. This can be achieved by sensing thedirection of current flow through the circuit breaker in order todetermine which of the two terminals is positive with respect to theother, so that the switches can be controlled accordingly. Preferablythe power supply also includes an arrangement to ensure proper start-upof the circuit breaker. In embodiments this comprises a reservoircapacitor charged by leakage current through the power switching devicewhen the power switching device is off In embodiments the power supplyfrom this leakage current is sufficient to operate a microcontroller orother circuit to sense the direction of current flow through the circuitbreaker, i.e. the orientation in which a circuit breaker is connected,before the power switching device has switched on, and thus when thedevice switches on can automatically start up the switched mode powerconverter to provide power of the correct polarity to the controller.

The invention further provides a circuit breaker operably connected tothe BJT structure above, the circuit breaker comprising: an inputcapacitor connected to a CE region; an inductor coupled to the inputcapacitor; first and second switching devices coupled to the inductor; asecond capacitor coupled to the second switching device; and a pulsewidth modulation (PWM) controller configured to control the first andsecond switching devices. It will be appreciated that the circuitbreaker can be operably connected to a standard BJT structure.

When a positive voltage is applied to the CE terminal, the firstswitching device may be configured to charge the inductor, and thesecond switching device may be configured to charge the secondcapacitor. The charging of the inductor may be controlled by controllingthe

duty cycles of the PWM controller. When a negative voltage is applied tothe first CE terminal, the second switching device and the secondcapacitor may be disconnected from the circuit breaker.

The circuit breaker above may further comprise a third switching deviceand a third capacitor which are coupled to the first switching device,the inductor and the first capacitor. The third switch may be configuredto charge the third capacitor.

The invention may further provide a bootstrap circuit operativelyconnected to the BJT structure above and operatively connected to thecircuit breaker above, the bootstrap circuit comprising a first diodecoupled with the second capacitor of the circuit breaker and a seconddiode coupled with the third capacitor of the circuit breaker, whereinthe bootstrap circuit is configured to store positive or negativeleakage current in the first and/or third capacitors through the firstand second diodes so as to turn on the bi-direction BJT structure. Itwill be appreciated that the bootstrap circuit breaker can be operablyconnected to a standard BJT structure.

The bootstrap circuit may further comprise a bleed resistor to providesufficient current to turn on the BJT structure if there is inherentleakage current present in the BJT structure.

The bootstrap circuit may further comprise an auxiliary tap circuitswitching on around the zero-crossing times so as to power the BJTstructure.

The invention may further provide a driver circuit operatively connectedto a plurality of BJT structures above, wherein each BJT structure isdisposed side by side on a chip and wherein the driver circuit comprisesa plurality of independent PWM drivers each independently driving thebase connection of each BJT structure through an inductor. Each PWMdriver may be configured to control current to the base connection andswitching time of the BJT structure independently. It will beappreciated that the driver circuit can be operably connected to aplurality of standard BJT structures.

Each PWM driver may be configured to control the current during anon-state of the BJT structure using a discontinuous current inductordrive.

The discontinuous current mode may occur when an off-time from the PWMdriver is sufficiently long so that the inductor current decreases tozero.

The invention may further provide a driver circuit operatively connectedto a BJT structures above or to a standard BJT structure, the drivercircuit comprising a resistive digital to analogue controller (DAC) forcontrolling the current of the base of the BJT structure. The DAC may beconfigured to control the base current of the BJT structure according toa control program which is reactive to measured operating conditions ofthe BJT structure.

The invention may further provide a matrix converter comprising an arrayof BJT structures above, the matrix converter further comprising acontrol circuit comprising a plurality of channels which are configuredto control the switching of the array of BJT structures.

The invention may further comprise a relay circuit for a low leakagecurrent application, the relay circuit comprising the BJT structureabove or a standard BJT structure, the relay circuit further comprisinga load resistor and a switching device arranged parallel to the loadresistor, wherein the switching device is configured to bypass anyleakage current from the BJT structure around the load resistor duringswitching off operation.

The relay circuit may further comprise a further switching devicecoupled with the load resistor, the further switching device beingconfigured to obtain Pico-ampere level leakage current into the loadresistor.

The invention may further provide a driver chip operatively connected toa BJT structure above or to a standard BJT structure and may comprisethe driver circuit above, wherein the driver chip is configured to applypre-programmed coefficients determined after manufacturing thecomponents of the driver chip.

The first PWM controller may be configured to vary phases for differentregions of the BJT structure based on calibration parameters of thedriver chip so as to allow a large die including the BJT structure toturn on and/or off to compensate for the difference in for examplecarrier lifetime and/or doping levels.

The driver chip and other circuit component including base inductors andstorage capacitors may be mounted directly on top of a wafer comprisingthe BJT structure.

According to a further aspect of the present invention, there isprovided a method of manufacturing a bipolar junction transistor (BJT)structure, the method comprising: forming a base region of a firstconductivity type, wherein said base region constitutes a drift regionof said structure; forming first and second collector/emitter (CE)regions, each of a second conductivity type adjacent opposite ends ofsaid base region, wherein said base region is lightly doped relative tosaid collector/emitter regions; and forming a base connection to saidbase region, wherein said base connection is within or adjacent to saidfirst collector/emitter region.

The method may further comprise: etching the first collector/emitterregion; and forming a diffusion region in the etched region. The methodmay further comprise filling polysilicon in a trench to form the firstcollector/emitter region and/or to form a thin interfacial oxide region.

The method may further comprise applying an anisotropic wet chemicaletching of the first collector/emitter region with artwork aligned ateither zero degrees or 45 degrees to form a simultaneous undercut of anoxide and a self-terminating V-groove etch of contact holes.

The method may further comprise applying the anisotropic wet etching toform a bevel etch to control the edges of the BJT structure. The methodmay further comprise applying an electric field grading technique toreduce minority carrier injection from the collector/emitter regions.The method may further comprise forming a three dimensional or stackedstructure so as to give higher power ability and/or higher sensitivityand lower conduction losses.

The method may further comprise forming a recessed BASE contact so thatthe electrodes on collector/emitter regions can form the threedimensional or stacked structure.

According to a further aspect of the invention, there is provided anactive rectifier comprising:

a power bipolar junction transistor (BJT), having a first and secondinput/output (I/O) connections and a base connection;

first and second rectifier terminals, wherein said first I/O connectionof said BJT is coupled to said first rectifier terminal, wherein saidsecond I/O connection of said BJT is coupled to said second rectifierterminal;

a driver oscillator to provide a two phase drive waveform having a first(on) portion and a second (oft) portion;

at least one controllable switch controlled by said driver oscillatorand coupled between said second rectifier terminal, said base connectionof said BJT and said second I/O connection of said BJT, to selectivelyroute current from said second rectifier terminal between said secondI/O connection of said BJT and said base connection of said BJT;

wherein said driver oscillator controls said controllable switch toroute said current from said second rectifier terminal between said baseand second I/O connections of said BJT in proportion of a ratio ofdurations of said first and second portions of said drive waveform.

The second I/O connection of said BJT may be coupled to said secondrectifier terminal via a filter, and wherein said filter may comprise acapacitor such that a connection between said second I/O connection ofsaid BJT and said second rectifier terminal is via said capacitor.

The active rectifier may further comprise an inductance between saidsecond rectifier terminal and said base connection said BJT to storecurrent for said base connection whilst said controllable switch isrouting current from said second rectifier terminal away from said baseconnection of said BJT.

The controllable switch may comprise a first controllable switch coupledbetween said second rectifier terminal and said second I/O connection ofsaid BJT and a second controllable switch coupled between said secondrectifier terminal and said base connection of said BJT and; and whereinthe two phase drive waveform may comprises first and second waveforms,said first waveform having an on portion corresponding to said firstportion of said two phase drive waveform, said second waveform having anoff portion corresponding to said second portion of said two phase drivewaveform, wherein said first waveform controls said first controllableswitch and said second waveform controls said second controllableswitch.

The controllable switch may comprise a first controllable switch coupledbetween said second rectifier terminal and said second I/O connection ofsaid BJT and a second controllable switch coupled between said secondrectifier terminal and said base connection of said BJT and; and whereinthe two phase drive waveform may comprise first and second waveforms,said first waveform having an off portion corresponding to said secondportion of said two phase drive waveform, said second waveform having anon portion corresponding to said first portion of said two phase drivewaveform, wherein said first waveform controls said first controllableswitch and said second waveform controls said second controllableswitch.

The active rectifier may further comprise a boost converter to boost avoltage drop across one or more circuit elements coupled between saidrectifier terminals to provide a power supply for said drive oscillator.

The boost converter may be coupled across one or more circuit elementscoupled in an emitter circuit of said BJT

The active rectifier may further comprise an inductance between saidsecond rectifier terminal and said base connection said BJT to storecurrent for said base connection whilst said controllable switch isrouting current from said second rectifier terminal away from said baseconnection of said BJT; and wherein said boost converter may comprisesaid inductance, to boost said voltage drop, and said driver oscillatorsuch that said driver oscillator, and inductance together with said atleast one controllable switch form a boost converter to power saiddriver oscillator.

The active rectifier may be configured to use leakage current throughsaid BJT, or a high voltage current source device, or a resistor, toprovide power to bootstrap said driver oscillator of said boosterconverter.

The on portions of said first and second waveforms are non-overlappingsuch that there is a dead time between said on portions; the activerectifier may further comprise a power harvesting device or Schottkydiode coupled to a connection between said second rectifier

terminal and said second I/O terminal of said BJT to harvest power fromsaid voltage drop during said dead time.

The first I/O connection of the BJT may be a collector connection andthe second I/O connection of said BJT may be an emitter connection.

The ratio of durations of the first portion to the second portion of thetwo phase drive waveform may be less than 1:1.

BRIEF DESCRIPTION OF THE EMBODIMENTS

These and other aspects of the invention will now be further described,by way of example only, with reference to the accompanying figures inwhich:

FIG. 1A illustrates an example dual-base version of a verticalcross-sectional structure of a double-gated device;

FIG. 1B illustrates a single-base version of a vertical cross-sectionalstructure of a double-gated device;

FIG. 1C illustrates an alternative single-base version of a verticalcross-sectional structure of a double-gated device;

FIG. 2A illustrates a drive circuit used for the devices of FIG. 1;

FIG. 2B illustrates a drive circuit used for the bi-directional BJTdevices of FIG. 1;

FIG. 2C illustrates a power scavenging circuit;

FIG. 3A illustrates hole current densities when operating as per FIGS. 1and 2;

FIG. 3B illustrates electron current densities when operating as perFIGS. 1 and 2;

FIG. 4A illustrates cross sections of an alternative BJT structure;

FIG. 4B illustrates a polysilicon emitter system of an alternative BJTstructure;

FIG. 4C illustrates an alternative BJT structure having a dual CE;

FIG. 5A is a schematic symbol of a BJT device which illustrates aP-channel JFET being effectively in series with the base terminal;

FIG. 5B is a schematic symbol of a BJT device which illustratesnon-encroachment of donor atoms in channel;

FIG. 5C is a schematic symbol of a BJT device which illustratesencroachment of donors in channel;

FIG. 6 illustrates a driver circuit;

FIG. 7 illustrates a concept view of an alternative transistorcomprising multiple parallel connected stripes and metallisationtogether with field-plate extensions for increased breakdown voltage;

FIG. 8A illustrates an array of chips in which the chips are inter-wiredusing a flex-pcb and wire-bonded to the individual die;

FIG. 8B illustrates the array of FIG. 8A folded;

FIG. 9 illustrates a layout of a 3D stacking of devices (folding) withfacility to increase surface area when even higher currents arerequired;

FIG. 10A shows an alternative route to a definite PNP input stage forthe BASE compared to the arrangement shown in FIG. 5;

FIG. 10B shows an alternative PNP input stage in which devices arestacked back-to back.

FIG. 11 is a 3D view of a minimal unit stack which can be scaled in X, Yand Z;

FIG. 12A illustrates structures which enable an either/or choice ofwireless/wired and has an additional advantage of furnishing power tothe attached device in wired-mode without having to break the wires whena node is attached to the network;

FIG. 12B shows a hinged magnetic transformer for use with the structuresof FIG. 12A;

FIG. 12C shows a terminator for use with the structures of FIG. 12A;

FIG. 13A is a schematic bootstrap/boost voltage circuit diagram whichshows a +Ve voltage conduction of the AC transistor;

FIG. 13B is a schematic bootstrap/boost voltage circuit diagram showingthe two time portions of the PWM cycle (1) and (2) feeding energy fromVCE1 to a total-loss circuit;

FIG. 13C is a schematic bootstrap/boost voltage circuit diagram which isoperational when I_LOAD is negative;

FIG. 14 illustrates process steps of a bi-directional BJT device(JFET-base transistor) using Nitride;

FIG. 15 illustrates process steps of a bi-directional BJT device(JFET-base transistor) using oxide only;

FIG. 16 illustrates process steps of a bi-directional BJT device(BJT-base transistor) using oxide only;

FIG. 17 illustrates the processing steps of manufacturing thebi-directional device (BJT base transistor) using a single mask in {100}and {110} etching methods;

FIG. 18 illustrates an alternative single mask scheme with self-limitingcontact depth;

FIG. 19 illustrates singulation/bevel/passivation steps for thebi-directional BJT device;

FIG. 20 illustrates electric field distributions in a bi-directionaldevice;

FIG. 21 illustrates the doping concentrations in a bi-directional BJTdevice;

FIG. 22A illustrates an array of CE1 stripes;

FIG. 22B shows an alternative view of the array of CE1 stripes in FIG.22A;

FIG. 23 illustrates a solid state relay module including a ‘slab·typeinductor for bootstrap DC-DC;

FIG. 24A is an illustration of a bi-directional BJT device (BJT PNPbase) in an off state with zero volts in terminals;

FIG. 24B shows the device of FIG. 24A in an off state having CE2 with apositive voltage and other terminals at zero volt;

FIG. 24C shows the device of FIG. 24A in an off state having CE2 with anegative voltage and other two terminals still at zero volt.

FIG. 24D shows the device of FIG. 24A in an on-state with CE2 at +0.1V,CE1 at 0V and BASE at +0.6V;

FIG. 24E shows the device of FIG. 24A in an on-state with CE2 at −0.1V.

FIG. 25A illustrates the off-state operations of an alternativebi-directional BJT device in which all terminals are in zero voltage;

FIG. 25B illustrates the off-state operations of the device of FIG. 25Ahaving CE2 with a positive voltage and other terminals at zero volt;

FIG. 25C illustrates the off-state operations of the device of FIG. 25Ahaving CE2 with a negative voltage and other two terminals still at zerovolt;

FIG. 25D illustrates the on-state operations of the device of FIG. 25Ahaving CE2 at +0.1V, CE1 at 0V and BASE at +0.6V;

FIG. 25E illustrates the on-state operations of the device of FIG. 25Ahaving CE2 at −0.1V.

FIG. 26A illustrates an arrangement of switches which operate in twophases during positive inductor charging phase;

FIG. 26B illustrates the arrangement of FIG. 26A during positiveinductor discharging phase;

FIG. 26C illustrates the arrangement of FIG. 26BA during negativeinductor charging phase;

FIG. 26D illustrates the arrangement of FIG. 26A during negativeinductor discharging phase;

FIG. 26E illustrates the arrangement of FIG. 26A initial bootstrapcircuit;

FIG. 26F illustrates the arrangement of FIG. 26A charge pump circuit;

FIG. 26G illustrates the arrangement of FIG. 26A base drive circuit;

FIG. 27A illustrates a driver circuit, specifically a multi-outputinductive base drive;

FIG. 27B illustrates example voltage waveforms for use in the drivercircuit of FIG. 27A;

FIG. 27C illustrates base pulse using pre-charge and discharge for usein the driver circuit of FIG. 27A;

FIG. 27D illustrates base finger driver for use with the driver circuitof FIG. 27A;

FIG. 27E illustrates alternative base finger driver to that shown inFIG. 27D;

FIG. 27F illustrates inductor driven base waveform for use in the drivercircuit of FIG. 27A;

FIG. 27G illustrates multiphase operation single-base connection;

FIG. 27H illustrates base on/off controlled by PWM 1.

FIG. 28 illustrates a transfer curve of current vs. PWM value (0-255range) for one path showing that discontinuous current drive is highlynon-linear;

FIG. 29 is a schematic diagram of a digital current mode driver;

FIG. 30A illustrates a cross section and equivalent circuit of astandard IGBT;

FIG. 30B illustrates a cross section and equivalent circuit of twoalternative BJTs;

FIG. 30C illustrates a cross section of an alternative BJT;

FIG. 30D illustrates a cross section and equivalent circuit of analternative IGBT;

FIG. 30E illustrates the doping profile of the device of FIG. 30D);

FIG. 30F illustrates Beta vs Current Density waveform;

FIG. 30G illustrates a top view and bottom view of a full die accordingto the devices above;

FIG. 31A illustrates main current path with a positive voltage;

FIG. 31B illustrates main current path with a negative voltage;

FIG. 32A shows a variable frequency matrix converter drive systemtopology for low cost and high reliability using the transistorstructures and driver techniques described in the previous embodimentsin which;

FIG. 32B illustrates triple driver module for the system of FIG. 32A;

FIG. 32C illustrates a boost circuit for the system of FIG. 32A.

FIG. 33A illustrates an example of a driver chip mounted to a powertransistor using an interposer flex-PCB;

FIG. 33B illustrates an example of a programmable PWM skew circuit;

FIG. 34 illustrates an example of a low leakage relay switch;

FIG. 35A illustrates an alternative scheme to a standard CMOS process tooptimise it for the role of driver especially of NPN versions of thepower transistor where most of the PWM conduction current is via NFETdevices to/from 0V.

FIG. 35B illustrates a simplified synchronous rectifier system forisolated power and data to/from driver IC using the CMOS chip;

FIG. 36 illustrates an example of a 3-phase inverter using DC bus andsynchronous mains rectification;

FIG. 37A illustrates an active diode concept;

FIG. 37B illustrates an example of 10 amp forward conduction (ignoringinductor and C 1 ripple current);

FIG. 37C illustrates an integrated version of a vertical B2 device;

FIG. 37D illustrates a hand-made demonstration of self-resonant circuit;

FIG. 37E illustrates an air-core or ferrite inductors;

FIG. 37F illustrates a Veroboard construction;

FIG. 38A illustrates a C2 device circuit;

FIG. 38B illustrates CMOS integration of an I2 device resulting in C2;

FIG. 38C illustrates drive waveforms;

FIG. 38D illustrates a plan view of a standard cell of C2 device;

FIG. 39A illustrates D2-control CMOS basis arrangement;

FIG. 39B illustrates D2-control for T2 transistor;

FIG. 39C illustrates B2 or T2 die having 3.3 mm.times.3.3 mm dimensions;

FIG. 39D illustrates a stacked die on printed-conductor substrate;

FIG. 39E illustrates an encapsulated bridge rectifier design;

FIG. 40 illustrates a metal assisted chemical etching process using amoving platform;

FIG. 41 illustrates a cross section of single sided, unidirectionalBMJFET structure;

FIG. 42 illustrates a B-TRAN device;

FIG. 43 illustrates a double sided Silicon Carbide IGBT;

FIG. 44 illustrates a bidirectional Bipolar Mode JFET and driver;

FIG. 45A illustrates a transient test circuit;

FIG. 45B illustrates symbols appropriate for Bipolar operation;

FIG. 45C illustrates a biplolar turn off sequence for high performanceswitching;

FIG. 46 illustrates TCAD falltime @ 100 A/cm2; and

FIG. 47 illustrates an Auxiliary Resonant Commutated Pole converter.

DETAILED DESCRIPTION OF THE EMBODIMENTS

An Example of a Bi-Directional Transistor Design

What follows is a general non-limiting explanation of the concepts andan initial design which may not be subject to well-known optimisationtechniques for highest gain, highest voltage withstand ability.

The present invention will be described with respect to certain drawingsbut the invention is not limited thereto but only by the claims. Thedrawings described are only schematic and are non-limiting. Each drawingmay not include all of the features of the invention and thereforeshould not necessarily be considered to be an embodiment of theinvention. In the drawings, the size of some of the elements may beexaggerated and not drawn to scale for illustrative purposes. Thedimensions and the relative dimensions do not correspond to actualreductions to practice of the invention.

Furthermore, the terms first, second, third and the like in thedescription and in the claims, are used for distinguishing betweensimilar elements and not necessarily for describing a sequence, eithertemporally, spatially, in ranking or in any other manner. It is to beunderstood that the terms so used are interchangeable under appropriatecircumstances and that operation is capable in other sequences thandescribed or illustrated herein.

Moreover, the terms top, bottom, over, under and the like in thedescription and the claims are used for descriptive purposes and notnecessarily for describing relative positions. It is to be understoodthat the terms so used are interchangeable under appropriatecircumstances and that operation is capable in other orientations thandescribed or illustrated herein.

It is to be noticed that the term “comprising”, used in the claims,should not be interpreted as being restricted to the means listedthereafter; it does not exclude other elements or steps. It is thus tobe interpreted as specifying the presence of the stated features,integers, steps or components as referred to, but does not preclude thepresence or addition of one or more other features, integers, steps orcomponents, or groups thereof. Thus, the scope of the expression “adevice comprising means A and B” should not be limited to devicesconsisting only of components A and B. It means that with respect to thepresent invention, the only relevant components of the device are A andB.

Similarly, it is to be noticed that the term “connected”, used in thedescription, should not be interpreted as being restricted to directconnections only. Thus, the scope of the expression “a device Aconnected to a device B” should not be limited to devices or systemswherein an output of device A is directly connected to an input ofdevice B. It means that there exists a path between an output of A andan input of B which may be a path including other devices or means.“Connected” may mean that two or more elements are either in directphysical or electrical contact, or that two or more elements are not indirect contact with each other but yet still co-operate or interact witheach other. For instance, wireless connectivity is contemplated.

Reference throughout this specification to “an embodiment” or “anaspect” means that a particular feature, structure or characteristicdescribed in connection with the embodiment or aspect is included in atleast one embodiment or aspect of the present invention. Thus,appearances of the phrases “in one embodiment”, “in an embodiment”, or“in an aspect” in various places throughout this specification are notnecessarily all referring to the same embodiment or aspect, but mayrefer to different embodiments or aspects. Furthermore, the particularfeatures, structures or characteristics of any embodiment or aspect ofthe invention may be combined in any suitable manner, as would beapparent to one of ordinary skill in the art from this disclosure, inone or more embodiments or aspects.

Similarly, it should be appreciated that in the description variousfeatures of the invention are sometimes grouped together in a singleembodiment, figure, or description thereof for the purpose ofstreamlining the disclosure and aiding in the understanding of one ormore of the various inventive aspects. This method of disclosure,however, is not to be interpreted as reflecting an intention that theclaimed invention requires more features than are expressly recited ineach claim. Moreover, the description of any individual drawing oraspect should not necessarily be considered to be an embodiment of theinvention. Rather, as the following claims reflect, inventive aspectslie in fewer than all features of a single foregoing disclosedembodiment. Thus, the claims following the detailed description arehereby expressly incorporated into this detailed description, with eachclaim standing on its own as a separate embodiment of this invention.

Furthermore, while some embodiments described herein include somefeatures included in other embodiments, combinations of features ofdifferent embodiments are meant to be within the scope of the invention,and form yet further embodiments, as will be understood by those skilledin the art. For example, in the following claims, any of the claimedembodiments can be used in any combination.

In the description provided herein, numerous specific details are setforth. However, it is understood that embodiments of the invention maybe practised without these specific details. In other instances,well-known methods, structures and techniques have not been shown indetail in order not to obscure an understanding of this description.

In the discussion of the invention, unless stated to the contrary, thedisclosure of alternative values for the upper or lower limit of thepermitted range of a parameter, coupled with an indication that one ofsaid values is more highly preferred than the other, is to be construedas an implied statement that each intermediate value of said parameter,lying between the more preferred and the less preferred of saidalternatives, is itself preferred to said less preferred value and alsoto each value lying between said less preferred value and saidintermediate value.

The use of the term “at least one” may mean only one in certaincircumstances. The use of the term “any” may mean “all” and/or “each” incertain circumstances.

The principles of the invention will now be described by a detaileddescription of at least one drawing relating to exemplary features. Itis clear that other arrangements can be configured according to theknowledge of persons skilled in the art without departing from theunderlying concept or technical teaching, the invention being limitedonly by the terms of the appended claims.

FIG. 41 shows a conventional single sided, unidirectional BMJFETstructure as per “The bipolar mode FET: a new power device combining FETwith BJT operation”, Microelectronics Journal Volume 24, Issues 1-2,January 1993, Pages 61-74, where the Source electrode goes to 0V powerand control, Drain goes to +Ve power potential, Gate is controlled +Veor −Ve with a small voltage with respect to Source.

To make the bidirectional version, this structure is duplicated on eachside of a N-type wafer—including as shown in FIG. 44 (only componentsinvolved in the shown switch conduction polarity are shown; fieldlimiting ring structures, common on most high voltage transistors arenot shown. In practice, a full complement of components are provided totop and bottom driver to allow bidirectional operation.)

It is an object of this invention that new circuitry as described willallow for this N-type-wafer double-sided BMJFET device to operate in amore efficient NPN operation than is possible for single sided BMJFET.

The double sided BMJFET characteristic can be well anticipated fromknown operation of the single-sided BMJFET where each side's gateelectrode would be driven to control a positive polarity existing on theopposite side of the device. Two separate drivers, one for each sidewould be needed. With the dual configuration there actually exists afundamentally better way to drive the device in bipolar mode.

FIG. 44 Shows the preferred operation where, unlike the classic BMJFETmode where the source electrode is driven from the low-side, it is heredriven from the high-side (whichever side happens to be high at thetime—the controller can determine this through CMP1 or CMP2).

The mode has been identified before in connection with a differenttransistor type:

In FIG. 2a an emitter-follower NPN mode can be invoked by driving thebase on the opposing side of the transistor to the side which is at thelowest potential. Because of the different connection system of thedouble-sided BMJFET described here versus that in FIG. 2a , thecorrespondence between terminal names as follows CE1 will be SOURCETOPand CE2 will be SOURCEBOT, BASE1 will be GATETOP, BASE2 will be GATEBOT.

The mode is an emitter-follower transistor configuration and notcommon-emitter and FIG. 31A and FIG. 31B give a numerical exampleshowing that there is a preferred mode of bipolar operation with reduceddevice current densities and where the base current is actually usefulin the load circuit and not entirely wasted in circulation. This appliesto standard BJTs operating in reverse direction, Bidirectional BJTs andBidirectional Bipolar-Mode JFET when operating in bipolar mode and withelectrode polarities as mentioned.

FIG. 44 has a diagram of a bidirectional BMJFET. Shown is an electricalsymbol representing the JFET characteristic of the device. The JFETcharacteristic keeps the device off when the gate is reverse biased andturns on in majority-carrier mode when the gate approaches zero bias.JFET mode is fast but suffers from a high “On” resistance (at best thisis Wafer_thk/Wafer_resistivity). This limitation was the original spurto the development of the BMJFET—to allow minority carrier injection byforward biasing the JFET gate/source diode junction (at which point GATEwould be referred to as BASE terminal)—just like a bipolar transistor.

UC2 is a microcontroller with PWM outputs able to drive the switchesSWA, SWB which are low on-resistance mosfets.

On the low side of the device, CMP2 is a voltage comparator. Either orall of these components can be integrated within IC1, IC2.

Note that only the components needed to illustrate operation in theshown Vsw polarity are present in the diagram. In practice, both topsideand bottomside have a full complement of components since the overallcircuit is to work as a bidirectional switch in either polarity.

SWA1, SWB1 are driven from a microcontroller (not shown) circuit on thehigh side identical to uC2 shown for the low side. The PWM ratio steersload current flowing through inductor L1 into either the P+ or the N+electrodes making an effective “Forced Beta” base drive current into theP+ base region i.e. the ratio of base current to emitter current (i.e.hole current: electron current in this configuration). By managing theswitch timing sequence, it is also possible to insert a non-conductionperiod and obtain a boosted VDD supply via D1, of the order ˜1V to ˜5Vtypically to power the microcontroller and other drive electronicswithout need for a external supply on the high side.

Diodes D2 and D3 represent internal body diodes of the large SWA, SWBmosfets and provide a default power path prior to boot-up.

On the low side, a copy of the same circuitry will be biased differentlybecause of the imposed switch voltage Vsw. Terminal SOURCEBOT will takeup a potential of perhaps 3V to 50V depending on the construction of theJFET region of the BMJFET and fairly independent of the Vsw. SOURCEBOTis effectively the source of an Nch JFET whose gate is GATEBOT.Conveniently, this voltage can be rectified and used to power-up the lowside microcontroller and other driver circuitry.

Control firmware can use the signal POS_DET to detect the correct modehighside or lowside to operate in. Highside operation was discussedabove. Lowside operation involves turning SWA2 and SWB2 on or off at thecorrect time to implement the best possible switching characteristic.

Although possible, it is not currently anticipated that the lowsideneeds to perform a high frequency, forced-beta type drive of it localsource/base.

Weak coupling between the inductors is indicative of the potential fortransmitting power or information between the highside and lowside.

Opto2 (and the not-shown Opto1) show a minimalist isolated on/offcontrol path of the switch from another control system.

FIG. 45 shows a Bipolar turn Off sequence for high performanceswitching. In FIG. 45a VBE1 represents the effect of control circuitSWA1, SWB1 forced beta system of FIG. 44 in producing a forward biasvoltage for the Gate/Base electrode. The diagram is drawn in such a wayto be compatible with entering into a transient TCAD model.

In FIG. 45b the electrical symbols are redrawn to show the bipolaraspects of the transistor giving three different ways to depict the sametransistor. It is important to realise that dependent on the terminalconnections, the same device can be operated as NPN or PNP.

These symbolic representations are used in FIG. 45c to show how thedevice is turned on as an NPN where conduction is highest by virtue of afavourable Electron vs. Hole mobility ratio. The sequence illustrationexplains how the device can be ‘rolled over’ from NPN to PNP conductionwhich has typically a lower beta and therefore is easier to turn offcleanly (an NPN turn off tends to oscillate with plasma generated andamplified—slowing down the turn off dramatically).

Even though PNP conduction is sub-optimal for conduction losses, theshort duration of this mode during turn-off has minimal impacts on theswitching efficiency.

Note that the DC paths for the switches include inductors L1/L2 of FIG.7. These inductors are in the order of milliohms of DC resistance so forthe purposes of FIG. 45c , they can be ignored.

FIG. 46 is an example turn-off waveform in TCAD simulation of a 240 uthick N− wafer BMJFET device driven in the way described and at 100A/cm2 current density. Fall time is around 33 nS for 1000V.

To reduce switching losses by the soft-switching method, Auxiliaryresonant commutated pole converter, the bidirectional auxiliary switches(depicted as A1, A2 in FIG. 47) can be replaced with a singlebidirectional BMJFET and drive circuit as described by this invention.

While emphasis has been given to bipolar-mode operation (minoritycarrier injection) of the devices, it should be pointed out that at lowcurrent operation, a fully JFET characteristic (majority carrier only)can be obtained by reducing the level of forward bias voltage to belowapproximately 0.6V for a silicon BMJFET which suppresses the bipolarmode.

An NPN type structure may be described but certainly a PNP structure isalso possible by reversing the doping systems. Where reference is madeto diffusion, ion implantation is also an option and so on.

FIG. 1(A) shows an example of the vertical cross-sectional structure ofa double-gated device. On each face of a lightly doped P− wafer arerelatively-deep diffused Nmed (doping ranging from 10.sup.18 cm·sup.−3to 10.sup.21 cm·sup.−3) regions to form the CE (Collector Emitter)regions 100. These Nmed regions are routed to metal contacts in theusual way via a highly doped N+ diffusion to make the CE electrodes 105.Metal contact to the P− wafer is achieved with a relatively shallow P+diffusion to make a Base electrode 110, 115.

Useful for working operation of the “Invisible Base” concept of JFETsecond transistor is the management of several design factors within acontrolled range. When the parameters are within a solution-space thenthe Nmed regions fully deplete the P− silicon around the Base contactarea and stop what would otherwise be a direct conduction path from theupper Base1 electrode 110 to the lower Base2 electrode 115.

The Base may be “invisible” because of this depletion but stillfunctions like a BJT Base when forward biased. The transistor may alsowork even when the P− base region is impinged by the N doping to adegree. In this case the operation of the base ‘channel’ may besomewhere between the physics of PNP and a Pch JFET.

Below are listed variables are given for an initial working solution fora 1200V device with a 50 mV on-voltage and a current gain of 15. Defaultdoping concentration profiles are assumed.

Silicon wafer thickness (120 micron)[O 150] dist_x_total_microns=50

dist_x_spacing_microns=1.O

dist_x_base_diffusion_microns=2.5

dist_y_shallowjunction_microns=1.O

dist_y_deepjunction_microns=5.0

substrate doping=1.5e14 cm−3

Nless doping=1e18 cm·sup.−3

N+ doping=1 e19 cm·sup.−3

P+ doping=1 e20 cm·sup.−3

The pitch of repeating patterns would be at “dist×total microns/2”intervals and metallisation would connect the multiple stripes togetherin the normal way to make a larger device. The width in the TOADsimulations was 10,000 microns.

FIG. 1 also shows the electrical symbol for the structures.

The dual-base structure of FIG. 1 (A) is intended to be driven bytransformer-coupled base windings. Minority carriers are injected on thetop and bottom of the structure through the P/N, i.e. Base/CE forwardbiased junction increasing conductivity between CE1 and CE2 terminals.The symmetry gives equal current gain (hFE) in both the forward andreverse polarity power conduction directions.

A single base structure of FIG. 1 (B) when driven according to FIG. 2a(but with Base2 terminal omitted) injects carriers only on the top sideand so hFe is approximately halved when not switching in the preferredCE voltage polarity quadrant. However, this structure has the bigadvantage of a single base drive circuit (directly coupled) and does notrequire masking and patterning on both sides of the wafer as is the casewith the alternative designs.

Finally, in one embodiment, the structure of FIG. 1 (C) is similar buthas two CE electrodes 130, 135 displaced laterally on the lower side ofthe silicon and relies on minority carriers being injected from ‘above’or the top side by the dedicated base and emitter region on the topside. This design may suit lower voltages and thinner silicon wafers.

In all cases above it will be appreciated that, in the bi-directionaltransistor design, the Base (i.e. the P of the NPN structure) is workingas the Drift region and supports the full ‘off voltage rather than theusual configuration of an NPN transistor where the collector (N regionof an NPN structure) would act as the voltage-supporting drift region.

All the structures described above have much lower losses and hencehigher efficiency than the standard IGBT device.

FIG. 3 shows the Hole and Electron current densities when operating asper FIGS. 1 (A) and 2 a.

FIG. 2b illustrates a drive circuit 200 used for the bi-directional BJTdevices of FIG. 1. The drive circuit 200 has a microprocessor control ofthe base current and can give complete safe-operating area, shortcircuit protection, zero-crossing on and off all defined by software. ASTM32F373 microcontroller for example is able to control up to 6switches at an added cost of $0.50 per switch. It has flash ROM for datalogging and UART I/O for communication.

In FIG. 2b , MOSFETs Q2 and Q3 drive a small inductor L using PWM outputand synchronous rectified buck-converter technique to efficiently createthe low (0.7V typical) base voltage and current to switch the maintransistor Q1 (or the bi-directional BJT) on. Q4 is a quick-turn-offdevice for Q1 and can impose a negative base bias which helps toincrease the breakdown voltage of Q1. Rsense1 and the ADC channels ADC4,ADC3 give feedback to the control program as to the instantaneous basecurrent and base voltage. Rsense2 with ADC2

measures the emitter current (which includes base current which can besubtracted out digitally). ADC 1 via the protection resistor Rprotectmeasures the transistor voltage drop (VCE1-CE2) when switched on. Analgorithm can adjust the PWM ratio until the targeted voltage drop(VCE1-CE2) is maintained. Just sufficient base current will be usedwhich can prevent deep saturation of the main transistor which otherwiserenders it slow to turn-off

ADC0 with voltage divider formed by Rdiv 1, Rdiv2 allows thezero-crossing time of the mains waveform to be detected (for optionalzero-crossing synchronised on/off of the power switch) and for thesmart-appliance metering application the total power delivered to theload is given by multiplying this value by the through collector/emittercurrent determined previously.

Description of Power Scavenging Concept (Illustrated in FIG. 2c )

A detailed circuit is described later but in principle thebi-directional BJT device can get its base current from the voltage dropacross itself while it is switched on providing that the Hfe (currentgain) is sufficiently high.

For example with 10 A passing through the switch, an assumed Hfe(current gain) of 20, and a VCE1-VCE2 drop of 0.15 volts and the Vbe toswitch the device on is 0.7 volts.

The power used in the base is 10 A/20*0.7 volts=0.35 W, while the powerloss over the switch is 10 A*0.15V=1.5 W. Extracting 0.35 W with anefficient DC-DC boost converter operating at the 0.15V requires ahigh-voltage switch Q1 1 which protects the low-voltage circuits whenthe main transistor is off For example, 2.333 amps is needed from this0.15V source to power the base. If Q1 1 itself is of BJT typeconstruction then it too needs a source of a lower base current from thescavenging power supply.

To make things easier and in pursuit of the lowest overall losses theintelligence of the microprocessor can be used so that Q1's VCE1-VCE2drop is deliberately higher while switched on during the low-currentportions of the mains cycle in order to extract and store energy intothe Vdd, Vss capacitors ready for implementing lower voltage drop in Q 1during the peak voltage times.

Description of the Charge Control Model:

With a microprocessor in control of the switching, and digital feedbackof all the analogue quantities in the circuit a charge-control model canbe executed in order to keep the power transistor switched on in themost efficient way without overdriving the base. Generally, a measure ofthe VCE1-VCE2 voltage drop is taken and if lower than a pre-set target,e.g. 0.1 volts, then more base current is commanded from the PWM.Knowing that internal charge is building up on the base/CE junctions andthe capacitance of these junctions and the required

minority charge needed to support a particular switched current, thequantity and duration of this base current boost can be regulated inorder to intercept the demand current through the device as it is seento rise. Similar algorithm can be used for reducing currents. Thealgorithm can also take into account the recombination lifetime to havea constant estimate of the charge available for conduction.

Flash Memory

With a self-writable flash memory in the microprocessor, each powertransistor can have permanently stored a calibration area setup aftermanufacture during test and referable when operational to improve theaccuracy of the algorithms and the reported measurements of the device.

Alternative Fabrication Technique

An example of the fabrication technique of the devices of FIGS. 1 and 2is described below.

Etched Trench:

From additional simulations it is found that an etched trench withvertical sidewall in the silicon, when diffused, gives a FET-‘base’structure of higher performance than a simple diffused planar junction.

It is also more economical in terms of time and equipment utilisation toetch a trench and make a shallow diffusion (.about.30 minutes total)than a deep thermal diffusion which can take many hours. It also allowsa sharper edge to the diffusion profile to be achieved.

[O 178] FIG. 4a shows the structure and the identified parts.

Another option is a “Polysilicon Emitter” system shown in FIG. 4b inwhich a heavily doped N+ polysilicon is used to fill the trench formingthe emitter. Similar N+ polysilicon emitter formed on the bottom sidealso. Polysilicon emitters have higher gain at higher current densitiesdue to the barrier for hole injection (from the base into the emitter)formed by an inherent oxide layer which forms between the silicon andpolysilicon.

Equivalent Circuits of the IBT Device (or the Bi-Directional BJTDevice).

FIG. 5 is a schematic symbol of the bi-directional BJT device.Equivalent circuits are shown for the structures discussed previously.In the first case (in FIG. 5(a)), a P-channel JFET is effectively inseries with the base terminal. This happens when the N+ does not diffuseall the way into the channel region under the base contact. In thesecond case (in FIG. 5(b)), where some donor atoms do make it into thechannel, operation of the base drive does not immediately fail.Operation becomes that of a lightly-doped base PNP transistor in serieswith the main P−

base region. Too much encroachment may reduce the operating efficiencyof the device and can slow down turn-off somewhat but it remainsoperational (in FIG. 5(c)).

Solid-State Relay Replacement (with Reference to FIG. 6)

FIG. 6 illustrates a driver circuit 600. The circuit 600 can serve asdrop-in replacement of a standard solid-state relay. The circuit may beable to operate from just 2 power terminals,

i.e. the contact terminals. The power available from the signal side ofthe relay (e.g. 5 mA@5 volts) is insufficient to power the IBT base eventhrough a transformer.

A power scavenging system is required in lieu of another source ofexternal power.

Referring to FIG. 6, when the IBT transistor is off, a high bleedresistance can tap a few micro-amps (which appears as slightly elevatedleakage current into the load) to power an ultra-low-powermicrocontroller circuit which can boot up and run at a low Khz-typefrequency.

A storage capacitor C on +VDC line has sufficient charge such than whenthe IBT is required to be turned on, there is enough energy to drive thebase at least initially.

With the IBT turned on there is now a low voltage (the voltage drop)across the contact terminals CE 1 and CE2 in proportion to the currentconducted multiplied-by the ‘On’ resistance of the transistor. Thisvoltage-drop is a parasitic effect but it can be used to extract asource of power for maintaining base current in the IBT. There are twoimmediately apparent options for this. First there is a special dual-CE2structure as shown in FIG. 4c . Secondly, an additional IBT operating atlower current density (and therefore low voltage-drop) can tap the CE2voltage and conduct it onto VTAP as an input to the DC-DC converter.

An example of option 1: Device terminal CE2Y will also have a voltage ofapproximately CE2X (since the P− region is filled with minorityconductive carriers). The voltage at CE2X can be boosted to give acontinuous DC power source for powering the base.

Depending on whether the voltage drop is +Ve or −Ve (detected by acomparator in the microprocessor) the microprocessor activates theswitches in the following logic sequence then repeats.

An example of option 2: If the main transistor has a gain of 25 atVCE1/CE2=0.1 V at 5 A/cm·sup.2 then it needs 0.2 amps/cm·sup.2@0.7V forthe base which is 1.4 A/cm·sup.2 from 0.1V. If equal area is given overto the tap IBT it will be operating at 3.times. lower current densityand proportionally lower voltage drop and higher HFE (gain) than themain transistor but still VTAP would likely be 70 mV instead of 100 mV.Iteratively solving this (and including the base current for the secondIBT) it is anticipated that the IBTs can be self-powered with a 2.times.increase in total device area and 1.5.times. higher voltage drop. Thevoltage drop is still around 10.times. lower than comparable switchingdevice on the market.

[OI 89] In both cases the circuit self-adjusts to the biasingconditions, for example that the current through the switch rises, thenso does the voltage-drop over IBT I, this gives more voltage availableto VTAP and more power to the base drive which in turn helps to lowerthe VCE1/CE2 saturation voltage.

+Ve voltage drop at CE2. Voltage needs to be boosted by .about.10.times.

[OI 90] Phase1. Agate=O, Bgate=I, Cgate=O, Dgate=1. Duration=IOO uSexample. [OI 91] Phase2. Agate=I, Bgate=O, Cgate=O, Dgate=1. Duration=10uS example.

[OI 92] [repeat] −Ve voltage drop at CE2. Voltage needs to be invertedand boosted by I I.times.

Phase1. Agate=O, Bgate=I, Cgate=O, Dgate=1. Duration=I I O uS example.

Phase2. Agate=O, Bgate=I, Cgate=I, Dgate=O. Duration=I O uS example.

[repeat]

For option 2, the bases of the two IBT transistors could be driven byindependently controlled PWM/Inductor circuits instead of both basesbeing driven together. This facilitates turn on of the Q2 independentlywhich could be used in addition to the bleed circuit to gather largercurrents via the load resistance by switching on around thezero-crossing times of the mains waveform. Low voltages could beefficiently gathered in this way and for many loads (e.g. heaters, largemotors) the small additional ‘leakage’ would not affect them.

Low Cost Manufacturing

[OI 97] One possible starting material is P− high-minority-lifetimemono-crystalline solar wafers which can be readily sourced in a range of50 u to 300 u thickness. The thickness and doping depend on chosenwithstand voltage of the device (thicker, lower doped for highervoltage). At current prices, processed P− silicon wafer panels for solarpanels having been through the following processing steps: Etch,Diffusion P and N, Contacting and Passivation retail for $0.02 percm·sup.2. Operating at 2.5 A/cm·sup.2 for very low voltage drop and highgain, I O cm·sup.2 of silicon (20 cm·sup.2 for self-powered) might beused for a 25 A device. The high-volume target silicon cost could be aslow as $0.50 c for a 25 A 1000V 0.1 W per amp loss SSR using thesetechniques.

[OI 98] A concept view of an IBT transistor comprising multiple parallelconnected stripes and metallisation together with field-plate extensionsfor increased breakdown voltage is illustrated in FIG. 7

In order to accommodate the large silicon areas required for lowestloss, a 3D stacking technique would be used. Since each slice is only0.2 mm thick, a 25 A SSR could be fitted into an area of 20 mm.times. IOmm.times.2 mm. No special additional heatsink (.about.$2 cost) would berequired.

FIG. 8 illustrates a chip in which the chips are inter-wired using aflex-pcb and wirebonds to the individual die. FIG. 9 illustrates alayout of a 3D stacking of devices (folding) with facility to increasesurface area in case even higher currents are required (inter-wiringomitted for clarity).

The package may be coated in encapsulant for environmental protection.

Minority Lifetime Choices

High minority carrier lifetime is important in the P− region to get highHFE (gain).

However, shorter lifetime could be a target for the N+ regions wherehigh doping also tends to kills HFE at low current levels—this help toincrease breakdown voltage.

Alternative Transistor Construction

FIG. 10 shows an alternative route to a definite PNP input stage for theBASE (re. FIG. 5). This scheme can have lower charge-removal during turnoff relative to JFET, but has some advantages in high-current density,and can allow a simpler 3D stacking system to increase current rating ina given footprint area. A typical process is given for an R&D lab. Theillustration is a minimal unit drawn as a half (right hand half)cross-section of a cylindrical symmetric design.

Firstly a P− doped wafer has 20 u deep Phosphorous diffusion on bothsides. This gives N+ doping levels near the surface and N− towards thebottom of the diffusion. A polysilicon layer or silicon-carbide heterojunction layer can be added on both sides to create polysilicon emittersfor higher gain if desired.

Then an oxide layer is grown on the top side of the wafer (or aninsulator layer is deposited)

Openings in the insulator layer are made using photoresist-expose-etch.

A silicon-etch (e.g. KOH) is performed to make a trench for the BASEelectrode in the places defined by the previous openings. The insulatoracts as a mask. Undercutting is helpful.

A boron diffusion is performed. The insulator acts as a diffusionblocking mask meaning that only a thin shell of P+ diffusion is made atthe walls of the trench.

Now a second set of openings are made in the oxide in the regions wherecontact to the N+ is to be made.

Aluminium is sputtered from above. This gives mutually-unconnectedmetallisation connection to the BASE and CE1 regions. Lower costalternatives such as metallic pastes could be used.

For higher current such devices can be stacked back-to-back as shown inFIG. 10b . Inserting metal sheet between each layer in the stack canobtain electrical connection. The stack could be pressed and fired tomake thermo-compression bonding.

FIG. 11 is a 3D view of a minimal unit stack which can be scaled in X, Yand Z. A BASE connection will contact to two silicon die using solder,conductive paste or thermo/compression bonding. Similarly the CE1 andCE2 copper sheets will each connect to two different die using perhapssintered silver powder or high temperature solder—on the CE1 sideprobably screen printed onto the copper plates in a pattern to match theCE1 openings.

The final connection scheme to the outside world is determined by theshape of the copper sheets which are stamped and bent to make a kind oflead-frame.

A thermo-compression process can ‘sinter’ the stack together attemperatures between 250.degree. C. and 400.degree. C. typically, orconventional high-lead solder can be used.

A final encapsulation with plastic (not shown) will allow a SMDelectrical mounting of the device onto a PCB.

Alternative Signalling Scheme to Communicate/Power the SSR Modules

Bluetooth and Bluetooth Low Energy are examples of communication systemfor short range RF data exchange. This system could be used as-is tonetwork multiple solid-state-relays (SSRs) of the type (describedpreviously in this document) by using for example a Bluetooth capableSOC including Microcontroller from, for example, Nordic SemiconductorInc or Texas Instruments as an upgrade for the system microcontrollermentioned previously. Normally an antenna at each end of such acommunications link permits wireless connectivity. These antennas can berealised from a specific PCB trace pattern whose layout is specific andtuned to the radio frequency involved—typically 2.4 GHz. For industrialcontrol networks wireless is not often used due to the possibility ofinterference and security issues. A wired network is used instead.

FIG. 12 illustrates structures which enable an either/or choice ofwireless/wired and has an additional advantage of furnishing power tothe attached device in wired-mode without having to break the wires whena node is attached to the network.

A twisted pair electrical wire 1210 (e.g. UTP or STP) is capable oftransmitting 2.4 GHz RF signals for lengths of 15 meters with acceptableattenuation. Underneath this RF frequency can be a low-frequency powerwaveform e.g. .about.20 KHz created by a central generator.

On an intelligent SSR node which is to tap this power and the RFcommunications, the UTP cable is locally ‘untwisted’ to produce a largerthan usual loop. This loop is placed inside a hinged magnetictransformer 1215 which is a type of planar transformer for the LF power

frequency and forms the primary. The hinge 1220 is then closed tocomplete the magnetic circuit. The secondary side of the transformer isa PCB trace on the SSR module and has terminals labelled X I and X2.Rectifiers and filter capacitors on the secondary permit extraction ofpower from power waveform.

This configuration allows for solder-free insertion and removal itemsinto the network and gives 100% electrically isolation. The ferrite usedfor the transformer needs to have high permeability to the LF and act asa transmission-line transformer for the RF.

If designed correctly, the PCB trace for the secondary of thetransformer is also able to couple the typically 2.4 GHz RF signal fromthe UTP cable loop and into a Bluetooth RF chip on the SSR pcb 1225.

An inductor-capacitor network and rectifiers can separate the LF powerfrom the RF frequencies.

Signals X I and X2 from the pcb-transformer become VDD, VSS and RF1,RF2.

With sufficient screening in the RF components and transformer, anetwork of many nodes would only be responsive to signals in the UTPcabling and not very sensitive to general external Bluetooth signals inthe building. This allows high reliability signalling and no possibilityof eavesdropping. It also means that bandwidth increases when runningmultiple independent RF-UTP networks which run without collisions.

At the end of the UTP network cable a terminator 1230 is added. Thisgives an LF return current path which effectively puts all the Nodes inseries to the AC power generator. Higher number of nodes requiresproportionally higher voltage LF power signal.

With the hinged PCB transformer open, the unit would operate in a normalBluetooth wireless mode which is good for low-security networks andduring development. In this mode power would have to come from aself-powered scheme as described previously.

Simulation of Bootstrap/Boost Circuit

FIG. 13 is a schematic bootstrap/boost voltage circuit diagram in whichthe current and voltages seen by the bootstrap/boost voltage circuitoperating on CE1 terminal of the AC transistor is illustrated. Thecircuit is based on that described in FIG. 6.

FIG. 13(a) shows a +Ve voltage conduction of the AC transistor. VCE1node has a capacitor C1 which is charged up by the I_LOAD currentpassing through the transistor. VCE1 works like a low voltage powersource. All the power is extracted through the inductor and passed ontothe BASE to replenish recombination and other losses in the base regionof the AC transistor.

FIG. 13(b) simplifies the circuit to show the two time portions of thePWM cycle (1) and (2) feeding energy from VCE1 to a total-loss circuit(which in practice is replaced by the power supply load the controlsystem—including microcontroller and the BASE drive). The arrangement ofFIG. 13(b) IS for operational when I_LOAD is positive. FIG. 13(c) isoperational when ILOAD IS negative which generally requires the voltageinversion necessary.

Closed loop regulation of the bootstrap/boost mode can be implemented bya digital algorithm operating with input ADC results from all therelevant voltages and current monitor points (of the kind shown inprevious diagrams) and outputting to appropriate PWM control of basedrive (similar to FIG. 2b ).

Appendix 1 lists a python-code simulation which works with both+Ve and−Ve loads

-   -   determined as per FIG. 6 (VTAP is same as VCE1 of FIG. 13). The        algorithm first locks up the loop to achieve a current balance.        When I_LOAD matches the average current in I_INDUCTOR then VCE1        voltage is stable (does not rise or fall).

An additional goal for the loop is regulation of the VCE1 to a targetvoltage. For example +0.1V or −0.1V is a good target. Given thisvoltage, the algorithm-locked DC-DC conversion process will extractpower to provide to the BASE a current of approximately 1 Watt per10-amps of I_LOAD current. Relating this to a typical VBASE of 0.75Vgives a base current 1.333 A and corresponds to a “forced-beta” of about7.5. The system automatically gives a base current in proportion to theload current without algorithm intervention but the VCE target voltagecan be algorithmically changed dynamically if the transistor is seen(through ADC using circuits explained previously) coming out ofsaturation.

Digital Fuse/Circuit Breaker

The current-limiting and self-bootstrap power allow the completeintelligent device to operate as a two-terminal device. The unit caninitially scavenge deliberate leakage current and periodically wake-upthe microcontroller as has been described previously. When switching to“On” conduction mode it can run from the boost/bootstrap power andmaintain very low total losses compared to a standard fuse. The boost involtage is sufficient to power normal control circuits even though themeasured voltage across the two-terminal ‘Fuse’ would be 100 mV or less.

Such a fully programmable digital fuse can stand in for a traditionalfuse in any application and has superior ‘clearing’ speed if needed orcan be programmed to emulate any type of slow, medium or fast fuse typewith a programmable ‘trip’ point. Obviously it has the advantage over astandard fuse in that it is not destroyed when activated and couldself-reset with time, or power cycling events etc.

TABLE -US-00001 APPENDIX I Source code for DCDC +/− converter simulator#!/usr/bin/env python # # - this one tries to close the loop # - can do+VE or −VE # with INVERTED_MODE # # In inverted mode simulates theproper mosfet sequences. # All currents are the right sign. #INVERTED_MODE : # Only thing that gets done is to VBOOST which remainspositive # and to cope with that, the switches are simulated. # theopposite sign of VBOOST is used in all the calculations # relating tohow it affects the change in currents. Plus the current into C2 getsswapped. L= 1Oe-6 IL=0.0 VCEI_NOM = 0.1 VBOOST_NOM = 3.3 VCE1 = 0.0 ;#-voltage drop to work off VBOOST = 3.3 ;#- running voltage on C2 VZENER =VBOOST RZENER = 0.5 fpwm = 30000.0 tcycle = 1.0 I fpwm ;#- cycle timetime=O #-work out initial pwm ratio for no net current accumulationPWM_ratio_NOM = 1.0                       - (VCEI _NOM/VBOOST_NOM) ;#-doesnt need the + 1 because current calculated later as ramp down isfrom difference between VCE andVBOOST    print    “PWM_ratio_NOM=”,PWM_ratio        NOM                      Iaverage_VBOOST=O  IL_peak_max=O IL_peak_min=O C 1=1OOOe-6 ;#- VCE I capacitor C2 =1OOe-6 I_CEI = 10.0;#- amps coming fromthe switch I_BASE=O.O ;#- actually the current running around the zenerin this case #-                  --------------------------------------------    def                  run_for_secs( seconds)           globaltime,IL,Iaverage _ VBOOST,IL_peak_min,IL_peak_max,PWJVI_ratio,tcycle,VCE 1,VBOO- ST,INVERTED_MODE tend = time + seconds while True : print “” ton = tcycle *PWM_ratio toff = tcycle - ton #Ramp up in current IL += ton * VCE I I L;#- current increased during the on time IL_peak_max = IL #- work out theaverage current during the buildup phase laverage_VBUILDUP =((IL_peak_max + IL_peak_min) I 2.0) * (ton I tcycle) ;#- with ton=O,this will be zero VCEI += (I_CE I - Iaverage_VBUILDUP) * tcycle I CI ;#-recalc capacitor voltage #if VCEI <O : VCEI =O.O #Ramp down in currentif INVERTED_MODE == True : #invert mode, ramp down rate given only byVBOOST (other side switched to gnd) IL -= toff * -VBOOST I L else :#normal mode, VCEI reduces the I ramp down speed because it subtractsfrom VBOOST IL -= toff * (VBOOST - VCEI) I L #NOTE, IL is allowed to gonegative --- using a switch, not a diode - it can pump current both waysIL_peak_min = IL                    !average_VBOOST = ((IL_peak_max +IL_peak_min) I 2.0) * (toff I tcycle) ;#- average current through theboot (diode) portion print PWM_ratio print “uS=”,int(time/l e- 6),“VCEl=”,VCEl,“VBOOST=”,VBOOST,    “toff                    nS=”,int(toff/le-9)          print “ IL_peak_max=”,IL_peak_max,“IL_peak_min=”,IL_peak_min, “I_CEI    ”,I_CEI  ,“!average_VBOOST=”,Iaverage_   VBOOST,“Iaverage_ VBUILDUP”,Iaver-age.sub.-- VBUILDUP if VBOOST > VZENER : I_zener_loss_on_C2 = (VBOOST -VZENER) I RZENER else : I_zener_loss_on_C2=0.0 ;#- no current if haventpassed zener threshold if INVERTED_MODE == True : VBOOST -=Iaverage_VBOOST * tcycle I C2 ;# simulate the switches and swap ba else: #normal VBOOST += !average_VBOOST * tcycle I C2 #add the resistivedrop VBOOST -= I_zener_loss_on_C2 * tcycle I C2; #in inverted mode, Idoesnt continue from C 1 when boosting, comes from GNU instead ifINVERTED_MODE == False : #Only for normal mode is VCEl affected duringthe boost time VCEl - !average_VBOOST * tcycle I C 1 ;#- the boostcurrent also comes out of C 1 time += tcycle ;# move time on if time >=tend : break #------------------- #gets the PWM which would bring               about a delta current change - based on currentconditions of C1 and C2 voltages (VCEl and VBOOST) # - positive deltacurrent is an mcrease current for the inductor # defcalculate_pwm_for_a_delta_I           (delta,      scaling)               #see          .med          globalL,tcycle,VBOOST,VCELINVERTED_MODE if VBOOST == 0.0 : VBOOST = 1 e-6 ;#avoid /0 #See .med for slightly different formula if INVERTED_MODE ==True : PWM_ratio_for_no_I_change = ( (0 * L) + ( tcycle * -VBOOST) ) I (tcycle * (-VBOOST + VCEl ) ) ;#-so can scale +/− around the zero pointPWM ratio cancelling = ( (delta * L) + ( tcycle * - VBOOST) ) I (tcycle * (- VBOOST + VCEl ) ) ;#- which would be usable if no scalingelse : PWM_ratio_for_no_I_change = ( (0 * L) + ( tcycle * VBOOST) -(tcycle * VCEl) ) I ( tcycle * VBOOST) ;#-so can scale +/− around thezero point PWM ratio cancelling = ( (delta * L) + ( tcycle * VBOOST) -(tcycle * VCEl) ) I ( tcycle * VBOOST) ;#- which would be usable if noscaling relative_PWM_from_nochange_position = PWM_ratio_cancelling -            PWM_ratio_for_no_I_change ;#- normalise   scaled_relative_PWM_from_nochange_positionrelative_PWM_from_nochange_position *              scaling    ;#-  scale retumval  scaled_relative_PWM_from_nochange_position +PWM_ratio_for_no_I_change ;#- then put    back again #clamp ifreturnval > 1.0 : returnval = 0.999999 if returnval < 0.0 : retumval =  0.000001 return returnval #------------------- #Iterate with variouscurrents {tilde over ( )}{tilde over ( )}{tilde over ( )}{tilde over( )}{tilde over ( )}#SET THE       TARGET    VOLTAGE  on        VCE 1VCEl _target   0.15;#-       can      be  +   orVCEl_tune_resistance_ohm = 0.2 ;#- effective if VCEl_target < 0.0 :INVERTED_MODE = True #VCE 1_target = 0.0 - VCEl_target ;#- Put it backto being a positive one else : INVERTED_MODE = False PWM_ratio =PWM_ratio_NOM old_VCEl = VCEl #Keep I_CEl as positive number, will useINVERTED_MODE (when VCE 1 is negative) to send currents the other wayfor I_CEl in [10] : if INVERTED_MODE == True : I_CEl = 0.0 - I_CEl forcycle in range (450) : #Can work out what the I imbalance in C 1 is byrate of rise of voltage on it delta_I = C 1 * (VCEl - old_VCEl ) Itcycle ;#- If new voltage greater than old voltage then cap is risignfrom not enough juice taken out #So work out a duty cycle which willsend towards zero #- see .med delta_I_gain = 0.75 #Think this anintegrating effect. Result gives the delta to change the current by ineach cycle based on the voltage error VERROR = VCEl - VCE 1_targetCHANGE = VERROR / VCE 1_tune_resistance_ohm ;#- if VCE 1 is too high,this gets positive print “ICHANGE”,ICHANGE PWM_ratio_cancellingcalculate_pwm_for_a_delta_I (delta_! +!CHANGE, delta_I_gain ) print“cycle#”,cycle print“delta_!”,  delta_!,  “PWM_ratio_cancelling”,PWM_ratio_cancelling  PWMratio PWM_ratio_cancelling old_VCEl = VCEl run_for_secs(tcycle) print“-------------------------------- -------------------------------------------------------------------------- ---------” print “- RESULT @ ICEl = ” I CEl print “ VCE 1 = ” VCE 1 print“------------------------------------------------------------------ ---------------------------------------- ---------”

Detailed Fabrication Techniques

The diagrams and notes are example methods of manufacturing thebi-directional BJT device.

The processes will mostly be with reference to a PNP input (main)transistor but the same techniques are applicable for the JFET version.

FIG. 14 illustrates the process steps of a bi-directional BJT device(with a JFET second transistor) using Nitride. These steps use acombination of Silicon Nitride and Silicon Dioxide as masking/insulatingmaterials. One advantage of these steps is that the lithography for twomasks can be done prior to the etching processes and the wafers do notneed to be returned to lithography part-way through production.

FIG. 15 illustrates the process steps of a bi-directional BJT device(with a JFET second transistor) using oxide only. As to the steps ofFIG. 14, these steps do not use Nitride but with the 2 oxidemasking/etch steps at different point in the manufacturing process.

FIG. 16 illustrates the process steps of a bi-directional BJT device(with a BJT second transistor) using oxide only. In these steps 2 oxidemasking/etch steps are needed

FIG. 17 illustrates the processing steps of manufacturing thebi-directional device (BJT base transistor) using a single mask in {100}and {110} etching methods. In these steps, one oxide masking/etch stepis used. This is advantageous since there is no need for a second masklayer and it also provides alignment accuracy. The process relies on theanisotropic wet chemical etching of crystalline silicon using KOH, NaOH,TMAH, EDP or similar solutions. The contact holes for CE 1 are smallopenings whose edges are controlled by the {111} planes to make 54degree inverted pyramids whose sidewalls can be calculated to come to apoint at a precise depth below the initial silicon surface. After this,etching stops which gives a

controlled depth independent of further increases of etch time. Thetrenches for BASE are oriented on the {100} direction and etch downwardsat the same rate as laterally and give vertical sidewalls with 100%undercut ration. Etching is not self-limiting and is controlled usingtime.

FIG. 18 illustrates an alternative single mask scheme with self-limitingcontact depth. In this scheme unrestricted trench depth usingcrystallographic anisotropic etch is used. This has the 2D mask artworkfor forming a PNP-BASE transistor 1805 on a standard {100} orientedsilicon wafer. The wafer flat {110 plane} is aligned to the bottom edgeof the layout. Provided that the CEI contact openings are small relativethan the desired trench depth then the CEI contacts will only penetrateinto the very heavily doped top region of the N+ CEI diffusion. A 3Dresult from an etching simulator is shown for KOH as the etchant. Fromthe cutaway view the underside of the etched pyramid profile can beseen.

The trench depth is not limited for CEI because the 2D outline forms aconvex shape bounded by {100} vertical planes (due to the 45 degreerotation of the artwork). This allows independent control of the trenchdepth determined by (etch-rate*time), and this BASE trench depth ischosen to cut into the phosphorous diffusion to a lower-doped regionwhere functionality of the PNP transistor is improved (see dopingprofile below). However, when Boron diffusion is applied to form theBASE of the composite device (i.e. the emitter of the PNP) it also dopesthe CEI contact openings to .about. 1 e20/cm3 with boron atoms. This maynot sufficient to over-compensate the previous Phosphorous diffusion of.about. 1 e21/cm3 so the contact remains ohmic and only the nature ofthe N+ is seen in CEI terminal. The next step would be to apply thermalAluminium evaporation where the oxide overhang (resulting from theundercut) makes independent contact to the CE I and BASE regions. If thegap between CEI stripes is minimised it might be possible for thealuminium to bridge over the small gaps in the oxide to form a singleCEI terminal with no additional wire-bonds.

FIG. 19 illustrates singulation/bevel/passivation steps for thebi-directional BJT device. To obtain a high breakdown voltage in asemiconductor material, it requires careful control of the edges of thedevice. At the edges and along the surfaces of the depletion regionleakage currents can arise which lead to much lower than expectedbreakdown voltage of the junctions. These issues have been solved m manyways in the past but for AC devices there are complications m that themethod 1 s likely to be symmetrical. Reference“double_bevel_edge_termination_for_bidirectional_devices.sub.—1974.pdf’is a paper from 1974 outlining the method of double-positive-bevel edgetermination where it was achieved using sandblasting. This same profilecan be achieved here with an anisotropic etch technique.

First the wafer is held by the front surface using a special vacuumchuck with rubber sealing strips. Then X and Y deep grooves are sawnwith a diamond-coated grinding wheel from the rear—almost the entire waythrough the wafer. This is followed by anisotropic etching again fromthe rear side of the wafer to yield the required double-positive-bevelprofile in about 20-50 minutes of etching with 15% KOH at 100.degree. C.Following the {111} planes up and down reveals a very smooth doublebevel angle which is ultimately intersected and terminated. The bevelexists around whole outside profile of the die, including corners. Tosmooth the facets, a final isotropic etch can be used. Etching stops atthe SiO.sub.2 layer. At the end of etching process, the devices are heldtogether by just a thin SiO.sub.2 oxide layer and an aluminium layerwhich are on the front-side of the wafer and can be easilysingulated—but first a passivation layer is needed to cover the sideprofiles of the device. This passivation seals out contaminants,controls the electrical charge on the layers and reduces surfacerecombination. For P-type silicon, use can be made of a commercial ofalumina (Al2O3) passivation technique from solar wafer production calledatomic-layer-deposition machines (ALD).

So-called spatial-ALD involves sending the wafers forwards and backwardsdown a line of alternative precursor gas chambers to rapidly build upmonolayers of material.

FIG. 20 illustrates electric field distributions in the bi-directionaldevice. The two lower pictures are the electric fields modelled withSilvaco Atlas tool of +1400V and −1400V applied between CEI and CE2terminals with BASE at 0 v. The simulation has cylindrical symmetryaround the X=O line. When the simulation is run without thedouble-positive bevel, breakdown voltage occurs below 900V due to theelectric field exceeding 2e5 V/cm at the surface of the silicon.

FIG. 21 illustrates the doping concentrations in the bi-directional BJTdevice. The doping concentrations are taken in the Y direction in a linestraight-through the wafer, missing the BASE region. The right-handtrace is the doping profile at the BASE region.

Base Resistors On-Die.

FIG. 22 illustrates an array 2200 of CEI stripes. On-die variations ofdoping levels, junction thicknesses and recombination lifetimes canprevent even a monolithic device acting as a single ideal device evenwhen the arrays of CEI stripes are hardwired together. Instead it willact as a parallel collection of slightly mismatched transistors. Thismismatching can result in current crowding in certain regions of thedevice, hot-spots and generally lower than calculated performance. Forthe BJT type devices operating with a voltage mode base-drive, a 2:1mismatch of base current between any two stripes can arise from about a25 mV (kT/Q) offset in locally-process-dependent BASE voltage of the twostripes.

To reduce this effect, series resistance can be added to the BASE ofeach sub-device in the parallel array. This has the effect of moreequally sharing the common BASE input drive current amongst thesub-devices despite VBASE differences.

The required series resistance can be obtained by use of the P+ diffusedregions which themselves are shallow and quite resistive. FIG. 22illustrates how an air-bridge can be created by defining a narrow sliverof SiO.sub.2 which upon etching is fully undercut and left suspended inmid-air.

SiO.sub.2 Air-bridge concept can also be used on the JFET-base device inorder to link up the multiple bases electrically to one another.

Slab Inductor for Bootstrap System.

It is known that adding bootstrap circuitry gets over one of the olderdrawbacks of BJT technology, namely that a not-insignificant currentdrive has to be found to drive the BASE of the device. Thyristors andTriacs although bipolar technology, actually self-power the BASEcurrents from the through-currents (inherent 0.7 to 1V drop however),but the end user does not have to create a continuous current drive.Therefore Thyristors and Triacs are still very popular for AC powerswitching applications.

FIG. 23 illustrates a solid state relay module 2300 including a ‘slab’type inductor for bootstrap DC-DC. It is effectively a I-turn ‘E’ core.The centre plate 2310 is twice the thickness of the top and bottomplates 2315, 2320. Flux is split 2-ways exactly like a standard E core.An air-gap default of the copper-foil thickness is apparent to allowhigh current operation without saturation.

The complete module 2300 is an intelligent power switch/relay/fuse ableto power itself from the through-current of the switching element whenon, and from leakage current when off

Bi-Directional BJT Device Theory of Operation and Drive Systems.

Operation of the different regions of the device follows the principlesof bipolar junction transistors and/or junction field effecttransistors. Primary conduction path is formed when minority-carrierinjection causes conductivity modulation of an otherwise lowly-doped,voltage-sustaining, “bulk” region. This path is similar to that in astandard Thyristor and is therefore well proven. In terms ofconductivity it is preferable to use a P-type semiconductor for the bulkwhere the minority carriers injected are electrons which have at least2.times. higher mobility and diffusivity versus holes. Another reasonfor using P-type semiconductors is that solar-cell P-type wafers, whichfeature an optimised high minority-carrier lifetime, are available atextremely low cost.

For the highest voltage operation (typically >2 kV) an N-typeconductivity-modulated region is commonly used due the availability ofnuclear-radiation doped silicon (radiation mutates silicon intophosphorous at a very well controlled rate).

A driver circuit is described later which supports both P-bulk andN-bulk devices and has facilities for bootstrap.

Description of the Operation:

FIG. 24 illustrates the off-state and on-state operations of thebi-directional BJT device in accordance with the present invention. Thediagrams attempt to show a coarse-grained doping level using circles torepresent electrons (filled) and holes (unfilled). The same symbols areused to show moving carriers during conduction. The diagrams show a halfof a device symmetrical about the X=O line. In practice many ‘stripes’are arranged in parallel to form a large device. All diagrams are forP-bulk devices. N-bulk devices would have all voltage polaritiesreversed and NIP doped regions reversed.

Off State Operation of a BJT PNP-Type Base Device

FIG. 24(a) is an illustration of a bi-directional BJT device (BJT PNPbase) with zero bias on all the terminals (CE1, CE2 and BASE). A smalldepletion region 2405 exists around each junction of the device.

FIG. 24(b) is an illustration of a bi-directional BJT device (BJT PNPbase) with a large positive voltage on CE2 with the other two terminalsstill at zero volts. A large depletion region exists between the CE2 andthe bulk, drift region.

FIG. 24(c) is an illustration of a bi-directional BJT device (BJT PNPbase) with a large negative voltage on CE2 with the other two terminalsstill at zero volts. The area of the sunken base region 2415 is muchsmaller than the CE 1 region 2420 with the N+ doping and this dopingcombined with the N− doping under the base helps to ensure that the P−bulk region 2425 is fully depleted giving much the same voltagebreakdown characteristic as a uniformly N++ top region would give.

On-State Operation of a BJT PNP-Type Base Device:

FIG. 24(d) is an illustration of a bi-directional BJT device (BJT PNPbase) with CE2 at

+0.1V, CE1 at 0V and BASE at +0.6V. The main current flows when thedevice turns on under a condition where CE2 terminal which is slightly+Ve (e.g. +0.1V). This represents the condition after fully switching ofa load has been achieved (prior to this there could have been over 1000volts over the switch) and now just an ‘ohmic’ characteristic is seenacross the switching terminals of the device. In the PNP base region2415, where the N part is part of the CE1 terminal, holes are injectedand diffuse through into the P-bulk region 2425 of the main

conduction NPN region making a low-voltage-drop connection between BASE2415 and P Bulk 2425. Although both CEI/P-bulk and CE2/P-bulk junctionsbecome forward biased, the larger forward bias exists to CE I/P-bulk(0.6V applied verses 0.5V applied over the CE2IP bulk) and so moreelectrons diffuse into the P-bulk from the CEI end than the CE2 end.Also, more holes drift to that end of the device to provide the ‘base’(recombination) current of the main NPN transistor.

FIG. 24(e) is an illustration of a bi-directional BJT device (BJT PNPbase) with CE2 at

−0.1V. This is generally the case after fully switching a negativevoltage to a load. In this case a nominal 0.6V Vbe is achieved with only0.5V of base voltage. The extra 0.1V is provided by the −0.1V CE2voltage. For the reasons given previously and as seen in FIG. 24(e),hole current flows more down the CE2 where electrons are being injectedmore rapidly.

Note on Doping:

For efficient operation of the integrated two-transistor system it isuseful that the built in voltage of the PNP base/emitter junction ishigher than the NPN base/emitter junction of the main input transistor.When this is done the PNP acts as a switch whose emitter hole currentmostly runs usefully to the bulk region 2425 instead of throughdiode-action to CEI. The built in voltage depends on the dopingaccording to the following equation.V builtin:=VtIn(NaNdNi2)  ##EQUOOOO1##

Here Ni is intrinsic carrier concentration at temperature and Vt=k*T/q.This is usually the case when the doping of the bulk region is very lowto support a high voltage but it sets a minimum doping requirement forthe base 2415 of the PNP (and hence a maximum etch depth of thisfeature).

Operation of a JFET-Type Base Device

It should be stated first of all that the JFET input circuit does notgive the device a high input impedance. This is because the JFEToperates in a ‘common-gate’ mode where the input terminal of thebi-directional (T2) device is effectively the source of the JFET.

Off-State Operation of a JFET-Type Base Device:

FIG. 25 (a) is an illustration of a bi-directional BJT device (JFET typebase) with a zero voltage condition. The N+ and N regions abutting theBASE will fully deplete the short vertical channel region 2510 of holes.Construction of this normally-off JFET type base normally requires verytight control of doping and geometry. Even the maximally doped N+ regionis only able to deplete around 3 microns of a typically doped P− bulkmaterial. This means that the vertical JFET channel 2510 will tend to beless than 6 microns in the X direction (assuming N+ is on both sides ofthe channel).

In FIG. 25 (b) the channel is still depleted when BASE=0V but in thiscase this may not have an effect since the high voltage drop issupported at the CE2 end 2520 of the device and there is no tendency forcurrent flow in the channel 2510 of the JFET type base.

FIG. 25 (c) shows that the CE1 and JFET depletion regions are joinedtogether. As the JFET region is only a very small area relative to CE1,there is negligible loss of ability to form a high voltage withstandingdepletion region in the usual way.

On-State Operation of a JFET-Type Base Device:

FIG. 25(d) and FIG. 25(e) show the on-state operation which is verysimilar to the PNP base type device as illustrated in FIG. 24 (d) and(e). The only difference here is that holes flow directly from the BASEto the bulk whenever BASE is higher than the pinch-off voltage of theJFET that was formed. Holes do not need to diffuse through a region ofopposite doping as was the case for the PNP-type base in FIG. 24. Anadvantage of the JFET type base over the PNP type base input is thatnone of the BASE current of the T2 device is lost to base current of thePNP input stage which saves about 5% of the input drive current. Afurther advantage comes from the fact that the pinch-off voltage atabout 0.3V is generally less than 1.times.Vbe and could allow a drivercircuit to remove charge from a saturated device without a reversebiased junction preventing it.

Improved Bootstrap Circuitry

FIG. 26 illustrates improved bootstrap circuitry m accordance with thepresent invention. Unlike a standard Thyristor or Triac, the T2 devicesdo not have a mechanism for self-sustaining turn on. For Thyristor orTriac this mechanism comes at the price of approximately 1 W of loss peramp of switched current. The T2 device can reduce this by a factor ofaround 10 by exploiting the inherent current-gain of the BJT structure.However this may use additional circuitry.

FIG. 26 A, B illustrate an arrangement of switches which operate in twophases to multiple a low voltage at CE1 of typically <+0.1V to VDDvoltage of typically +1V. The MOSFET switches (Q1, Q2) are controlled bya PW1YI controller within a microcontroller. Q1 switches to charge theinductor 2610 with flux with C 1 acting as input filter capacitor. Q2switches the positive boost current to VDD and onto C2. Voltage boost isset by the duty cycle.

FIG. 26 C, D illustrate arrangements in which CE1 is negative (as itwould be when an AC load current reverses). The current build-up in theinductor is of the opposite polarity. When Q1 disconnects in this case,Q3 is switched to VEE to generate a negative voltage of typically

−1V. A different switch Q3 steers the negative boost to C3. There is astandby mode where this DC-DC converter can be shutdown to save power.

FIG. 26 F) is a self-explanatory charge pump circuit. This canbi-directionally couple

+VDD to −VEE transferring energy from one rail to the other under themicrocontroller command. This means that +1V and −1V (2V total) isavailable to power the microcontroller whether or not the T2 device isswitching is positive, negative or AC.

Drive Circuit Improvement

FIG. 26 G) is a base drive circuit which works with VDD, VEE and GND andis again driven by a PWM within the micro-controlled chip. This can becontrolled to produce a positive base current for a NPN type T2 device(P-bulk region) or a negative base current which would drive a PNP typeT2 device (N-bulk region). In discontinuous mode, the PWM ratio sets theBASE current.

At 100% PWM the maximum voltage available to the base is either +VDD or−VEE.

Initial Bootstrap Circuit

Prior to the Inductive DC-DC switch on, and prior to the switch on ofthe charge pump the circuit must accumulate enough energy to be able tosuccessfully start-up. FIG. 26 E) illustrates a circuit which can takeeither positive or negative leakage current and store it on VDD (C2) orVEE(C3) via the diodes D1 or D2. Circuitry inside the control chip e.g.low frequency microcontroller mode, can start up on just 1.2V fromeither VDD or VEE.

Providing that C2 and/or C3 are large enough there would be sufficientenergy for a self-sustaining boot up of DC-DC and charge pumps when theT2 device is commanded to turn on. Until a command to turn on thedevice, an ultra-low power standby mode can be maintained which suppliedonly from leakage. If the inherent T2 leakage current (thermallygenerated semiconductor leakage) is present, then a bleed resistor inparallel (shown) can ensure sufficient current. Q9 is an NMOS switchwhich is able to isolate capacitor C 1 during the start up, since C 1might be of such a high value that during 60 Hz AC operation its voltagedoes not swing sufficiently to allow the diodes D 1, D2 to operate.

Controller Chip Construction

Many of the MOS switches used in bootstrapping operate at high current(same current as going through the T2 device) but because they are lowvoltage switches they are fabricated on the same die using the samedeep-sub-micron transistors as the microcontroller. “On” resistance forNMOS FETs built on a 0.15 u process are approximately 0.0005Ohm*mm·sup.2. It therefore only needs several mm·sup.2 of die area toimplement those

MOS Switches.

T2 Devices as being a Replacement of IGBTs

BACKGROUND

This document expands on the driver circuits for the T2 device(previously called IBT in the text—particularly but not exclusively tothe JFET input transistor device). It notes that a re-think is neededfor the role of current-mode transistors in modern electronics. It alsolooks at how to make a functional equivalent to the ubiquitous IGBTtransistor at less cost and with better performance.

Comparing Current-Mode to Voltage-Mode Drive of Power Semiconductors inGeneral: Voltage Mode Gate Drive:

MOSFETs and IGBTs have a voltage-mode on/off control where a voltagesufficiently high (10V typ.) on an insulated-gate causes an inversionchannel in the underlying lowly-doped semiconductor switching it fromoff to on. Once this voltage has been established on the gate then nomore current is needed to maintain the switch in the on-state. Intheory, the driver circuit need only be a low current circuit and issimple to make. In practice, when switching quickly, many amps ofcurrent are needed to allow fast charging of the gate for efficient turnon and off of the device. The average gate power consumption is stillvery low but even so, MOS devices require a high voltage (typ. 15Vrated) control driver process able to supply several amps of pulsedcurrent. Historically, when the power MOSFET and IGBTs appeared in the1970s, 1980s most control systems used +/−15V supplies so the highvoltage gate was not seen as a problem. Today it is rare to see analoguecontrol circuits operating above 5 volts and logic ICs operate at lessthan 3.3V.

Current-Mode Base Drive:

The bipolar junction transistor (BJT) is the archetypal current-modedevice which requires a current drive into its base to turn it on. Thevoltage level is low, generally less than 0.9V for a silicon device overtemperature. Initially the current goes to charge the junctioncapacitance but continuous current flow is required thereafter tomaintain the switch in the on condition. The continuous base currentresupplies the injected carriers which are lost due to recombination inthe device. The higher the ‘Beta’ current-gain of the transistor, thelower the continuous base current needs to be.

Base current power loss is often seen as non-negligible, yet it isgenerally overlooked that this reaps disproportionate gains inefficiency elsewhere in the device. For example a 25 A collector currentBJT with a Beta of 10, needs 2.5 A of base current to stay fully on.Assuming a base voltage of 0.75V this equates to 1.9 W of power. To makea fair comparison between various MOS and BJT type devices, (and in thebase of bootstrap base-current generation where this base power is notseen externally), it is useful to consider this loss as an

effective addition to the ‘On’ resistance of the devices. The exampleloss just mentioned equates to 3 mOhms of on resistance which for ahigh-voltage MOSFET is an impossibly low figure to match.

For efficient operation, a BJT device should be driven with‘just-enough’ base current to maintain the collector current at low Vee(sat). With the technology available in the 1970s, base drive circuitsto achieve dynamic control of saturation were bulky, expensive andnon-optimal. Therefore BJTs were quickly displaced by power MOSFETs atleast in the low voltage application area.

Today however where a single chip microcontroller is able to fullycontrol a BJT in real-time (32-bit microprocessor including FLASH, RAM,12-bit ADC, Quad PWM, Serial ports can be bought for .about.$0.50) thesituation is completely different.

Furthermore, low voltage, high current systems such are now common—atypical server microprocessor is powered over PCB traces at 150 amps at1 volt, and inductor based DCDC modules to supply these needs are small(recently this entire system including inductors are integrated on-chipon the Intel Haswell microprocessor).

Advantages of Current-Mode Low Voltage Drive

can be implemented in standard logic process with PWM, adc and drivers(low voltage

.about.0.9V maximum needed for a P/N junction in silicon)

possible to integrate the with microprocessor with driver forsystem-on-chip

reverse base current can turn off devices quicker than without (compareto IGBT)

Adiabatic recovery of a much of base charge (inductor versions)

Inductive Base Drive Circuits for Current-Mode Power Transistors

Previously the benefits of having multiple base current balancingresistors have been discussed to ensure matching of the BJT fingers (seethe description with reference to FIG. 22). Should there be a problemwith uniformity of silicon characteristics within the die of a singledevice. The same balancing effect can be achieved by modification of thepreviously described inductive buck-mode PWM driver to split the outputinto multiple inductance branches of the base current driver. This isshown in FIG. 27a for matching on-die but could also be used to matchmultiple die which are operated in parallel. Again this feature isoptional. An advantage of using inductors to match currents is thatthere are lower losses than using resistors. The variation of currentfrom one device to the next is proportional to the difference betweenVBE and the peak drive voltage to the inductor.

During the ‘On’ period, the average base current can be controlled usinga discontinuous-current inductor drive. Base current is proportional tothe PWM ratio 2.

Discontinuous operation occurs when the off-time is long enough for theinductor current to drop to zero before the next positive pulse. Thediscontinuous nature will not cause switch-off of the T2 device if thePWM frequency is sufficiently high. When the PWM period is short theamount of charge given in the PWM period is much less than thesaturation charge accumulated in the BJT junctions of the T2 devicewhich act as a capacitive filter to the base current.

FIG. 28 is a transfer curve of current vs. PWM value (0-255 range) forone path showing that discontinuous current drive is highly non-linear.

Advantageously, if the PWM inductive paths are split and independent, itgives an opportunity to add multiphase operation.

FIG. 27b suggests that the same PWM duty cycle is given to each of thepaths or advantageously the waveform of each path may also bephase-offset between each other to help to reduce switching noise andreduce ripple current.

Using ADCs, the microcontroller can be aware of and command the positiveor negative current build-up (turn-on, turn-off respectively) by holdingthe output positive or negative until the desired current is calculatedor measured to exist. Or it can set a level which is good for continuousconduction—just sufficient base current to keep the voltage drop overthe device at the ideal point based on saturation detection ADC resultfrom the CE2.

Fast Turn on Scheme with Inductors:

One downside of series-inductive base drive is that it limits therate-of-rise of base current (by definition of inductance) and thismight increase the turn-on time of the power transistor and thereforeits switching losses. For most applications, and with say a 1 MHz+PWMfrequency and low value base inductors (<1 uH), the fact that the mainT2 junctions will not turn on strongly and immediately (until a P/Njunction charge has been established) lets the base inductive currentbuild up strongly so that it ultimately pushes through the loss-dominantturn-on stage very quickly (less than 2 uS). If higher turn-on speedsare desired, then the IC transistor driver topology shown in FIG. 27A or27F can sidestep the normal delay in base current build-up using apre-charge scheme. Since the driver IC is intelligent and digital, andthe time of the positive drive current is known in advance, a pre chargeperiod can be deployed to get the inductor current up to the requiredlevel before applying the current to the base.

FIG. 27 C has the turn-on scheme waveforms. The current is applied tothe base only after it has fully ramped to the desired level.

Turn Off:

Turn-off can be done using opposite-voltage opposite-current drive.Storage time (time before all the charge is removed with the transistorstill conducting well) allows the current build-up without the need forspecial measures as used to turn it on. But, a more complex drive schemecould be envisioned where the inductor is electrically ‘swapped’ [DPDTMOS switches] to instantly reverse the base current keeping itsmagnitude and this would afford a very quick turn off.

Alternatively, an intelligent driver is able to reduce base currentahead of the known time that the transistor is to be switched off,letting it come out of saturation, and this will speed the transistorspassage through the turn off region reducing toff losses.

Adiabaticity:

During turn-on and turn off much of the charge transferred to the baseis adiabatically stored/recovered by the inductor/supply capacitorsystems. This allows use of very large +ve,

−ve currents to be deployed during turn-on, turn-off of the devicegiving high switching speed and lowest switching losses.

Capacitive Speed Up of on/Off:

Of course capacitor-base conventional BJT drive circuits could be used.

Fastest Digital on/Off Current-Mode Driver for High Speed IGBT-TypeApplications

FIG. 29 is a schematic diagram of a digital current mode driver 2900.Multiples of these can drive base fingers or die. The example is for anN-bulk T2 device such as the NIGBT structure (described later) where themain transistor is PNP and the base is either an NPN or N-JFET needing a−Ve polarity current to turn on.

The power source VEE_(adj) typically −0.9V comes from a DC-DC convertermade using a single inductor in the normal way (could be the bootstrapDC-DC) and is programmable by the VDAC set sufficient to drive the Vbeof the highest-voltage finger plus some margin to account for FETvoltage drop. Feedback from the ADC information can be used to fine tunethis voltage which could be set higher just prior to turn-on of the T2device for speedup.

This scheme is non-adiabatic but the current build-up does not depend oninductor ramp-times.

Fully digital ‘DAC’ control of individual BASE current outputs.Resistance between the BASE (VBE) of the device and the slightly greaterVEE (adj) (typically a 0.1 to 0.2V difference) sets the BASE current

Digital size-weighted FETs form an array (3 bits shown, could be 8 ormore) making effective binary setting of this resistance (formed of theFET ‘on’ resistance) connecting

VEE_(adj) to the BASE of the T2 device finger(s) and gives an overallDAC function for current.

Multiple units of the above can independently drive multiple basefingers and/or multiple die.

Turn-off of the T2 device is by the JFETs in this example. Unlike astandard IGBT it is possible with the T2 device (especially the JFETversion which has .about.0.3V barrier potential) to actively removecharge from within the junctions and the bulk of the device during turnoff Advantageously this shortens the turn-off time and reducesswitch-off energy loss.

Because a low-voltage, current-mode base drive is used, the system canbe implemented fully within in any standard deep submicron CMOS ASIC. Toget an idea of the area needed, a 0.18 u CMOS process (1.8 volt) hasNFET on resistance of less than 0.001 ohm*mm2. For example, a 100 A basecurrent driver requires only 1 mm·sup.2 of silicon NFET area in the ASICto switch this current (in practice getting the current in and out ofthe die takes up more space as does the buffers but the cost is stilllow).

Digital control permits ‘on-the-fly’ adjustment of drive at multiplepoints in the output power cycle. For example, at turn-on a very highcurrent could be given and during the cycle the base current can beregularly trimmed to keep the T2 transistor at the optimum point ofsaturation. Close to the end of the on-period, the current could bereduced to minimise the stored charge prior to finally an oppositecurrent used to turn the device off

A more advanced version could programmatically trim out mismatches inthe various characteristics of large transistors through pre-programmedoffset and gain currents between finger drivers determined bycalibration. This would give more uniform current density in the T2devices.

It may be more desirable to mount the ASIC on top of the T2 die and wirebond between the die.

Combination Drivers

One or more of the approaches mentioned can be combined for optimumeffect.

It might be possible to integrate some or all of the control electronicsonto the T2 transistor die.

I2 Device: A Combination of T2 Device and Driver Replaces IGBT with aNIGBT “Non Insulated-Gate-Bipolar-Transistor

Review of Structure and Operation of the Standard Planar IGBT

FIG. 30 (a) illustrates a cross-section of a standard planar IGBT. Whenthe gate terminal is taken to typical +10V, an inversion sheet forms onthe P region under the gate giving an N−

type connection from the N+ emitter to the N− base region. This featuredoes not in itself inject minority-carriers which instead are injectedfrom the collector P+ region when the collector is positive. This hasthree sub-optimal effects. 1) The On-state voltage can never be lessthan 1 diode drop. 2) Although the device is otherwise capable of ACvoltage blocking, it cannot be switched on for the reverse polaritydirection. 3) There is no externally available contact to the N-regionwhich might be used to remove charge to effect a rapid switch-off

The equivalent circuit for an IGBT is often represented as a PNPtransistor with its base switched by a NMOS transistor. This is a goodmodel for those devices where the PNP beta is designed to be relativelyhigh (>4). In those devices the ‘base’ current flows as electronsthrough the NMOS while a much larger current of holes diffuses from C toE as minority carriers in the base.

Another model that of a PIN diode in series with an NMOS devices isappropriate where the beta of the PNP<.about.2. In these cases,recombination is so high that most injected holes do not make it from Cto E but are present in enough numbers to reduce the resistance of theN− layer like a PIN diode. Current in the NMOS is a higher proportion ofthe total switched current in this case with less aid from thetransistor Beta action.

Structure of the 12 Devices Used to Replace IGBT

A N− bulk version of a 12 device see FIG. 30B made with either a BJTbase (in this case an NPN base) or a JFET base I2 device each operate ina similar way by injecting electrons into the N− base and are justreverse-doped versions of I2 structures already described. Polaritiesand current flow directions are reversed relative to a P− bulk T2device.

Whereas in an IGBT, J-FET regions are considered ‘parasitic’ and lead tothe development on the trench IGBT to avoid them; in the I2 JFET-basedevice the JFET region is encouraged and essential for the off-state. Inthe on-state the depleted channel disappears to be replaced with aminority-carrier-injected channel orders of magnitude higherconductivity than a :NIOS inversion channel.

Operationally, the main current-carrying PNP transistor is virtuallyidentical to the IGBT PNP mode (and for that matter to that of aThyristor), but in this case it can operate without a built-in diodedrop in the C to E (CE 1 to CE2) main conduction path.

The gate/base arrangement is very different. All of the insulating gatematerial and contact of an IGBT are no longer present and there is no N+region connecting to the E any longer. This eliminates the possibilityof NPNP latch-up because the N+ contact is not to a potential orpermanent low impedance capable of sustaining latch-up.

For these N− bulk I2 devices, a negative base voltage of −VBE at currentof Ibase gives BETA.times.Ibase of current through C to E terminals. Incases of low BETA of the PNP, the PIN action will also be present (likea PIN-mode IGBT) and the base will have to take most of the switchingcurrent of the device. In the limit, with if there is no PNP action, theefficiency of the I2 device will not much exceed the IGBT except for theelimination of the MOS resistive channel voltage drop of an IGBT.

Punch-Through/Non-Punch-Through/Electron Irradiation Options:

The same set of design optimisations used for IGBTs such aspunch-through/N+ buffer layer can be applied to the I2 devices making itasymmetric and not able to support (much) reverse voltage.

The region label BL can be N−, N+ or something in between, exactly likestandard IGBT processing for Field-Stop, Soft-Punch-Through,Controlled-Punch-Through, Light-Punch through etc. Electron irradiationcan be applied to the T2 type devices as it can to traditional IGBTs forlifetime control.

Transparent Emitters/Collectors:

There also seems to be no reason that a transparent emitter or collectorcould not be achieved also if desired. This occurs where the doping isso shallow that the carriers can pass right through and recombine on thehigh-surface recombination velocity metal contacts.

Inherent Inverse Diode Possibility

For any version of the I2 device the AC conduction ability of thestructure can give ‘for free’ a programmable inverse-parallel diodefeature (whereas the IGBT has no way of turning on the thick-base BJTwhen the collector goes negative). When the I2 driver IC detects anegative collector voltage it can optionally drive the base current inwhich case holes will be pulled out of the top P+ junction turning onthe transistor and clamping the negative excursion. This clamping actionis ohmic (saturating) so can be as low as 0.1V at moderate currentdensities.

I2 “NIGBT” Advantages Over IGBT

much simpler fabrication process (does not need oxide, fewer masks, lessdiffusions).

much more efficient

free of the PIN diode drop

‘ohmic’ like saturation ‘on’ resistance, or non-saturated under programcontrol.

Direct control of the internal charge of the main base junctions

Possibility to do fast switch off by actively removing charge from thebase

AC or DC switching

inherent anti-parallel diode action (ohmic) with special driver (whereasIGBT needs an external diode or extra processing steps to fabricate one)

Example Conversion of IGBT to NIGBT I2 Device

To illustrate the process of converting an IGBT to a 12 device, we tookthe planar IGBT example distributed with Silvaco's Atlas TCADdrift-diffusion simulator.

See http://www.silvaco.com/examples/tcad/section40/example4/ for theexample.

With the same doping and geometry, stripping out the MOS part andrearranging the terminals, and contacts, an 12 version reduced thesimulated on-voltage from 1.75V to 0.15V at the cost of (equivalent VCEloss) 0.5V due to base current. This cost dropped to 0.2V equivalentwhen the lifetime was increased from 1 uS to 5 uS (still well withinstandard CZ wafer specifications).

The overall benefit is a 2.5-5.times. reduction of conduction losses,improvement in switching performance, a driver that can integrated on aCMOS chip, and 2 fewer manufacturing masking steps.

AC Current Conduction Paths

When considering the AC operation of an I2 device, FIG. 31 helps tovisualise the main current paths in each direction of operation. In thiscircuit, the Beta of the device overall device is 10.

(Note: Beta=Alpha/(1−Alpha); Alpha=Beta/(1+Beta))

The 12 device can be further simplified for a DC application in whichcase it does not require a second transistor element such as a JFET. SeeFIG. 30 C in which a lateral version of the 12 device is illustrated).The construction techniques mentioned before are usable but the etchedbase feature could be deep enough to cut right through the CEIphosphorous diffusion.

Matrix Converter Application

Possibly the single largest application for medium voltage, low/mediumfrequency power switching is in variable-frequency motor driveapplications—currently $18 Bn per year for these inverter drives.

Roughly 50% of all electrical power used in the world goes into electricmotor applications where a variable frequency drive can reduce this byaround 18% typically (sourcehttp:!/en.wikipedia.org/wiki/Variable-frequency_drive).

Currently, motor applications which have been converted tovariable-frequency operation are claimed to save over 1 billion tonnesof carbon per year in emissions.

Variable-frequency drives represent 3% of the installed base and areonly fitted to 40% of new motors entering the market (the rest aredirectly connected to 3 phase line power) leaving a huge untappedpotential carbon saving.

The biggest reason to the low adoption rate is cost of the inverterelectronics and secondary problems caused by increased input harmonicpower from standard switching electronics such as IGBT AC-DC-AC baseddrives and reduced reliability due to DC bus capacitor failure.

The well-known and extensively researched Matrix Converter AC-ACconverter can circumvent all the known issues and has the followingadvantages for motor drive applications.

Very low input harmonics

Good power factor

High reliability by exclusion of DC capacitors

Bi-directional energy flow—breaking, regeneration possible (Matrixconverter topology is actually a general AC-AC power conversion topologynot restricted to motor driving).

FIG. 32 shows a Matrix converter system topology for very low cost andhigh reliability using 12 devices and driver techniques already outlinedin this document. An array of 3.times.3 AC switches is fundamental tothe operation of direct Matrix converters and the I2 devices aresuitable for this application.

When grouped in arrays of 3.times.12-devices they can be driven locallyby 3-output driver IC containing 3 copies of any of the various drivecircuits discussed. A central control circuit can coordinate theswitching of the 3 banks of triple-devices.

Device Naming Conventions:

T2 device—non-punch-through design with almost symmetrical forward andreverse conducting and voltage withstand ability. It has a 2 transistorstructure and also has high voltage, high current main BJT powerstructure—NPN or PNP. In addition, T2 devices have an input basetransistor of either JFET or BJT construction, for example, T2 NJ—an NPNwith a JFET input transistor, and T2 PB—an PNP with a bipolar inputtransistor. It will be appreciated that the T2 devices are previouslyreferred to as an IBT device, particularly but not exclusively, designedfor an IBT JFET transistor version.

I2 device—punch-through, or field-stop device to compete with IGBT andalso known as “NIGBT” (non-insulated gate bipolar transistor). Thisdevice is the same as the T2 but the addition of a field-stop or otherburied layer reduces the voltage withstand ability in one direction toaround −20V while the current capability of both directions ispreserved. The I2 device have a two transistor structure—same options asT2 (see above). I2 devices operate in

“Emitter follower” configurations and are useful even when the maintransistor Beta drops below unity.

B2 device—usually but not necessarily punch-through, or field-stopdevice to compete with IGBT so another kind of “NIGBT”. Like I2 devicethe voltage withstand ability is low but now—0.6V max. 1 transistorstructure—a vertical BJT. When operated with correct driver it has aninherent reverse diode which comes ‘for free’ and can be conductivitymodulated by reverse saturation by the base for very-low forwardvoltage. B2 devices operate in common-emitter mode.

Field-Stop (FS) Devices.

A Field-stop is no different in principle to a punch-through designdiscussed before where there is a buried layer of extra-high dopingwithin the voltage sustaining base (drift) region of the device.

(See FIG. 30, where BL is the buried layer). The field-stop is oftenjust thicker and less highly doped than a buried layer but to the samepurpose. Referring to an NPN (P-base) device, when the reverse bias ofthe high voltage junction exceeds a certain limit, the depletion regionfully extends through the P-base layer 3015 and only the higher P-dopedField-stop layer 3010 prevents the depletion region from reaching theEmitter N+ diffusion 3025 and causing breakdown. The device has a muchhigher breakdown voltage for a given die thickness in its preferreddirection but is no longer a symmetrical AC switch, since in the reversedirection the field stop layer's doping has the effect of dropping thebreakdown voltage considerably. Nevertheless DC switches are preferredin many applications.

The effect of the field-stop on the T2, 12 and B2 devices is similar tothat in an IGBT or PIN diode where punch-through operation also helps tospeed up the device turn-off without needing additional lifetime controlmeasures.

FIG. 30D illustrates a schematic diagram of a B2 device including avertical thick-base BJT with a Field-Stop layer 3010. This design is afully planar version but anisotropic-etch versions are also possible. Ahalf-section of 20 u width is shown.

The field-stop layer 3010 for an NPN B2 device will be a P diffusion oftypically 9 u thick and doping levels of around 1 e15 to 1e16/cm·sup.3can be formed with either Boron, Aluminium or Gallium impurities, thelatter two having fast diffusion speed and can possibly be co-doped(co-fired) with the phosphorous impurities for a single furnacediffusion process yielding the desired junctions. A suitable referencefor this process can be U.S. Pat. No. 3,681,155A.

SiO.sub.2 can be used as the mask material against phosphorous to formthe junction patterns.

FIG. 30 G illustrates a top view and bottom view of a full die madeaccording to the scheme (before metallisation). Field rings on thebottom side are for edge-termination and reduce the electric fields atthe border of the die (unlike an IGBT, in the B2 device, the depletionregion starts at the collector (bottom) 3020 side).

Because the device operates in common-emitter mode, the Beta graph ofFIG. 30F is particularly important because at Beta less than.about.O.75, the device becomes less efficient than an IGBT. At theintended operating point beta is around 2.5 with an effective Vee (sat)effective of .about.0.5 volts and even lower at lower current densities.

Driver IC Improvements

It will be noted that the description here refers to multiple basefingers of a single power transistor die but applies just as well tofingers spread over multiple power transistor die. It assumes alwaysthat there is a microcontroller or other sophisticated digital machineacting through a control algorithm to make PWM and other outputs basedon information received from ADC channels (See for example FIG. 2B).Also PMOS transistors are used for positive switching which could besubstituted by NMOS with the appropriate bootstrap or auxiliary powercircuits. The drivers here are useful for all kind of current-inputdevices such as Silicon Carbide BJT, GTO Thyrisors, Standard BJT devicesand BMJFETs (bipolar-mode JFETs)

Inductive Base Drivers

FIG. 27 A, B, C illustrate drivers which are able to drive NPN or PNPdevices because of a +/− output stage;

FIG. 27 D, E are base finger drivers which can also be built withstandard CMOS low cost process but simplified for one polarity drive orthe other, not both at the same time;

FIG. 27 F graphs the Spice simulation of the drivers and indicates thedesign features needed of the PWM mechanism to implement a flexible,efficient and very fast base drive response.

Dual PWM Per Base-Finger Driver

The key idea for a fast on/off current at the base is the dual-PWMsystem where the PWM1 is implemented with a fast low voltage on-chiptransistor (since it only sees Vbe) and gates the inductive current intothe base. It can also rapidly discharge the stored charge of the base.

PWM1 is just like a standard type PWM control signal for the transistor(although the PWM positions can be controlled in a specific way andindependent from one base finger to another as will be outlined below).

PWM2 is a faster-rate PWM which controls a synchronous buck-typearrangement on the inductor L1 which converts from a higher DC powersupply voltage to the Vbe voltage.

As indicated in the waveform, PWM2 can be used to pre-charge theinductor current within PWM1's off-time before it is gated onto the Baseoutputs at high speed and full strength.

During the ‘On’ period, the PWM2 can by dynamically adjusted e.g. fordynamic saturation control based on the instrumented readings of thetransistors operating conditions from ADC channels like FIG. 2B. Keepingthe transistor just within saturation prior to complete switch-off givesthe lowest E.sub.off (energy off) figure.

FIG. 27 D, E have ADC take offs at the base finger itself and from acurrent sense resistor ahead of the base. This permits a measurement ofactual base current and base voltage.

Independent/Semi-Independent:Multi-Finger PWM Drivers

Each finger-driver has its own PWM1 and PW:NI2 generator driven undersoftware control to control the On-Time, Off-Time independently of theother finger drivers. This ability can be used in software correct fornon-uniformity over large power-transistor die.

FIG. 33(B) shows a logic circuit which is able to re-time an existingPWM signal from say an existing PWM control IC. By tuning the final onand off position independently any mismatch between transistor regionscan be tuned out (mentioned before) but also any transistor Storage-timedelay, which increases the off time, distorting the PWM signal, can behidden with a corresponding extra delay to the turn-on position—thefinal switching waveform will be a delayed but overall faithfulreproduction of the PWM input. Most analogue feedback control ICs willnot notice this slight delay. Combining Multi-Phase Drivers for RippleReduction on a Single Output:

FIG. 27 (G) gives another other option for multi-phase multi-finger basecurrent generators feeding into one terminal—good for standardtransistors e.g. silicon-carbide BJT devices. Multi-phase operation ofeach driver can smooth out the base current ripple as can be seen fromFIG. 27 (H) while still maintains the very fast On/Off base currentcontrol. It has the net effect of a higher PWM2 frequency butmaintaining the low losses of the lower frequency. 3DDriver/Power-Transistor Stack:

The physical construction shown in FIG. 33A which has a driver chippermanently attached to a power transistor (T2, I2, B2 or other) usingan interposer flex-PCB. Inductors, capacitors and other surface mountcomponents can be added. An intelligent driver IC is solder

mounted to the PCB from above flip-chip style (WLCSP (Wafer Level ChipScale Packaging)) using bumping technique. The power-transistor deviceis solder mounted from underneath again using wafer-scale bumpingtechnique.

Other options could see the interposer made from a silicon substrate toeliminate any CTE (coefficient of thermal expansion) problems betweenthe die. Or, the main transistor die can have formed upon it additionalpatterned dielectric (e.g. polyimide)/metallisation layers toeffectively build an equivalent of the interposer directly on the powertransistor where the inductors and driver die could then be soldered.

On-Die Temperature Sensing of the Power Transistor at Multiple Sites:

Each base finger represents a PN junction and as such has a well definedforward-voltage (Vf) vs. temperature. Rather than taking the absolutevoltage measurement of Vbe which is one option, it is more accurate totake a Vf measurement at one current level and then another measurementat a different current level. The difference in readings varies withtemperature and eliminates a large unknown initial Vf voltage.

The inductor-driven base waveform (FIG. 27F) has within it varyingcurrent levels and so by sampling at intervals where base current ishigh and again where base current is lower using a different ADC channelfor each finger driven then a reasonable map of die temperatures isalways available to the control software. This is without needing to addspecific temperature sensors to the system or driver die (anotherpossibility).

FIG. 35 (A) is a small modification to a standard CMOS process tooptimise it for the role of driver especially of NPN versions of thepower transistor where most of the PWM conduction current is via NFETdevices to/from 0V. The N+ source diffusions have extensions deeper thannormal to connect directly to an N+ substrate which in turn can be

solder bonded to the metal lid of FIG. 33.

The process modification is small—similar to a deep N-well procedurecommon on CMOS processes.

The resulting NMOS devices have fully vertical current conductionpaths—between substrate and the bond pads. This means that no lateralcurrent is taken through the normally thin metallisation of CMOS chipsso avoiding the issue of electro-migration which tends to limit currentto a few mA/micron width. Having the bond pads over the active siliconarea means they do not add to the active area consumed.

FIG. 35 (B) illustrates a simplified synchronous rectifier system forisolated power and data to/from driver IC using the CMOS chip. Phasemodulation of the edges of the AC

switching by first the master (Driver) then the Slave (CMOS driver chip)can pass digital information in each direction along with powertransfer.

Software and Firmware:

When the features described above are combined with a microcontrollerwith on-chip non-volatile (NV) memory there exists an opportunity tofinally fix several problems which have plagued minority-carrier devicessuch as GTO Thyristors and BJT devices in the past.

These problems include:—

Current crowding on large die leading to premature failure of thoseparts of the transistor

with higher gain, higher temperature or higher carrier lifetime than themean.

Part-to-part variation of current gain—where replacement parts do notfunction identically to the original.

Turn-off, dynamic breakdown hot-spots—leading to failures in GTOThyristors.

After manufacture of the Driver/Transistor stack, a calibration routinecan be performed on the complete device.

One such test the Open-Circuit Diode-test which gives an approximationto the minority carrier lifetime and can be done even without aspecialised test fixture. The driver can turn on a Vbe junction andsimply measure the rate of droop of voltage with the Vbe junction afterletting it float.

Results of the calibration are stored in non-volatile memory on thedriver IC e.g. Flash/OTP or fuse memory.

Calibration routines can performed in a dedicated test fixturesequentially exercising each of the finger driver channel and determineperformance of only that area of the power transistor die. As well ascalibration data, the non-volatile memory can hold defaults such astemperature trip points.

The calibration values are later used during normal operation of thedevice to compensate for the inherent variations within the powertransistor and external components (e.g. inductance variations) so thatfrom the end-users view every device operates correctly and nearlyidentically to any other device of the same type.

For example, if the driver chip software accepts a digital input forcurrent limit in mA it can use its own lookups of Beta scale factors ofeach finger driver for the X and Y region of the device (with 2D lookuptable), and by controlling On time and Off time at independently of eachPWM finger driver channel to meet the commanded current.

Current crowding on a large die will be reduced and inter-changeabilityis realised with easy series/parallel connections of devices.

Parameters which vary with temperature can be tuned out given thetemperature measurement from the particular region of the die and knownparametric variation of transistor parameters with temperature.

Timing variations over large die are known to cause dynamic breakdownfailures of power devices during turning off especially those which relyon long minority-carrier lifetime to maintain low switching losses. Forexample GTO Thyristors (PNP wide-base transistor at the core) haverelatively good turn-on characteristics but during switch off,variations of carrier-lifetime, resistivity and doping levels over thedie area during dynamic breakdown makes the turn-off process uneven.This can mean that finally a small region which happens to be last toswitch off must momentarily conducts all of the switched current and canburn out. This problem can be solved with the smart driver usingindependent pre-calibration timings applied via PWM for eachfinger-driver to offset for the turn-off time variations.

Finally with a digital interface there is the opportunity to hide theunavoidable Storage Time parameter of BJT type devices where it takes aset amount of time to remove the charge from the base before the devicecan turn off. If the smart-driver accepts PWM turn-on and turn off timesas digital words, the already-known storage-time can be subtracted fromthe given turn-off time to position the turn-off edge at the correctpoint in time.

Matrix Converter Boost System

The AC version of the T2 device seems ideal for Matrix converterapplications compared to standard DC-only switches. Only one bigdrawback remains for the Matrix converter motor drive in that it canonly create an output of approximately 86% of the input voltage. Thisstops the matrix converter working as a drop-in replacement for standardDC-bus 3-phase inverters.

A simple solution for this would be to boost the AC input voltage by 14%using auto transformers.

However with 50 or 60 Hz transformers these would be bulky and eliminatemost of the advantages of the matrix converter. To address thislimitation a high-frequency AC-AC power converter is proposed in FIG.32. Unlike an AC→DC-AC based high frequency transformer, it has inherentbi-directionality so it does not lose the inherent advantage of a matrixconverter with regards to braking and regeneration.

The AC-AC transformers are not large being only 14% of the power of thematrix inverter.

Conventional 3-Phase Motor Inverter

T2, I2 and B2 type devices can still be applied to reducing power lossesand cost in standard AC-DC-AC topologies like FIG. 36. Here the devicescan perform mains-voltage rectification and output switching ateffective forward voltages of around 0.25V reducing the power loss by afactor of around 4. The I2 device is particularly suited as a high-sideswitch and a single driver can activate many switches with a commonsupply rail—unlike NFET or IGBT high-side switching systems whichrequire separate floating supply and driver chip for each driver.

Low Leakage Relay Contact

Compared to electromechanical relays, the on-resistance of the T2switches can be made equally low by increasing the device area. Leakagecurrent can increase in proportion to the area of the device and evenwhen switched off there could milliamps of leakage—enough to causeproblems for some loads.

FIG. 34 proposes a solution using a shunt element Q2 device to bypassthis leakage current around the load when QI is switched off and whereQI is much lower on-resistance than the load resistance this solutionwill work well.

To take the leakage current to Pico-amp levels even with very lowimpedance loads, Q3 can be added and it turns on and off with QI. Whenon it has to take the full load current but when off only has to supportunder IV typically and so could in fact be built onto the driver chipwhere ultra-low on resistance FETs occupy only a few mm·sup.2 (1.5mOhm*mm in 0.35 u CMOS).

Active Self-Powered Switching Devices and Circuits

These embodiments relate primarily but not exclusively to the driving ofBipolar Junction Transistor devices coupled to control circuits whichmay use BJT, MOS or other FETs. Descriptions here are generally furtherembodiments of ideas described in the embodiments above and especiallyrelevant background materials are FIG. 2C, FIG. 6, FIG. 13A, B, C, FIG.26, FIG. 27, FIG. 30 B,D. FIG. 33A, and FIG. 36, which are describedabove and which are applied often in relation to Circuit Breakers.

BJT devices require a Base current which is a fixed proportion of theCollector current to maintain saturated (low loss, ohmic) conductionmode. In this saturated conduction mode minority-carriers conduct bydiffusing from the Emitter to the Collector terminal with the Basesupplying carriers lost to recombination and at potential sufficient toovercome the built-in PN junction potential—typically 0.7V in silicon,and up to 3V for high-band-gap semiconductors such as Silicon-Carbide.

For a BJT transistor whose typical characteristic “Beta” [ratio ofcollector current achieved vs base current given] is 20, a circuitdesign point of 10:1 Collector:Base operating current might be chosen.This gives sufficient extra base current to account for part-to-partvariations and ensures the transistor will be fully saturated to a lowVCE (sat) voltage when turned on. In the example just given it is deemedthat the transistor is operating with a “Forced Beta” of 10.

In the literature there are very many application circuits to give aforced-beta for a transistor but generally these are not dynamic. If aBJT base is driven at fixed 1 A but the collector load varies from 1 Ato 10 A, only at 10 A collector-current is the forced-beta equal to 10.At lower loads the forced-beta is lower and power is being wasted in thebase (=Ibase*.about.0.7V). If the load happens to exceed 10 A thetransistor might be liable to turn off (insufficient base current) andcould easily overheat as it VCE rises dramatically.

Historically, BJT devices began to fall out of favour in the 1990s, notbecause of the switching efficiency—which is unmatched amongsthigh-voltage devices—but because of the cost involved in providing a(relatively) high base current which has to dynamically adjust inproportion to the instantaneous collector current. This provedproblematic, especially compared to power MOSFET and IGBT devices whichhave no such requirements.

It is one object of the present embodiment to provide a simple andautomatic means of dynamic base drive of BJT-style devices.

One device of the BJT era which is still prevalent in the market is theThyristor. This device ‘scavenges’ Base currents (recombination losses)directly from the current in the circuit in which it operates and doesnot need dynamic base control. In a Thyristor the full load currentpasses through a PNPN junction stack, or equivalently a PNP/NPNcross-coupled-and-merged transistor (the NP junctions of the PNP aremerged with the NP junctions of the NPN transistor). The Base currentsare provided naturally from the larger flow of full current carriersthrough the device. Unfortunately the silicon thyristor must have avoltage drop of at least 0.7V in the full load-current path to overcomethe remaining PN junction voltage and this voltage drop is at the fullload current. Also once the thyristor is turned on, it is difficult toturn off since the base currents come directly from the through-current.

It is another object of the present embodiment to furnish base drivecurrent from the circuit current but overcome the usual 0.7V voltagedrop and a further goal to make non-latching devices possible.

More recent prior art devices which seek to ‘scavenge’ power from theload to maintain their turn-on are based on MOS technology which haveessentially zero gate current making it

easy to use a charge pump or similar to extract operational gate voltagefrom across the terminals of the ‘closed’ switch. This approach howeverhas three drawbacks:

It requires a disconnect-circuit to decouple the driver from thehigh-voltage side of the switch at switch-off time.

The present embodiment aims to eliminate this by using only low-voltage(<5 volt) devices and capacitor operating on the low-voltage side of themain transistor with no need to touch the high voltage side of the maintransistor at all.

Extracting driver power from a boosted version of the voltage over the‘closed’ power switch means that the switch can never be fully turned-onfor if it were, there would be no operating voltage with which thedriver circuit can operate.

The present embodiment theoretically allows the main switch to turn onto zero-volts and no control needs to be applied to maintain aparticular switch voltage drop.

MOSFET devices are very large, inefficient and/or expensive whendesigned for high voltage operation.

Bipolar high-voltage device technology is the only option to compete oncost in most applications.

Embodiments Diode

D2-Device

The features of the method are described by way of an example, thesimplest embodiment being that of an active rectifier. Externally thisis a two terminal device and acts as a low-forward-voltage, high reverseblocking voltage diode. Four such devices can form a bridge-rectifier.

FIG. 37 shows the circuit and structures for producing a 2-terminal‘diode’ device with a forward voltage of the order of 0.1V and a reverseblocking voltage of >1000V at very low leakage current. This performancecontrasts with traditional high voltage silicon rectifier diodes whichexhibit a forward voltage between 0.75V to 1V and cause a significantpower loss especially on 120 Vac mains rectification applications (2diodes conducting) of around 2 W per amp which equates in monetary termsto $2 per amp, per annum wasted on electricity per bridge-rectifier.

Silicon Schottky-type rectifiers are not useful at high reverse voltagebecause a) the reverse leakage current is very high and b) a Schottkydevice designed for high reverse voltage unavoidably results in a highforward voltage.

FIG. 37B shows the operating principle given in context of a ‘diode’ butequally applicable when driving true transistors such as an I2 device(as described in relation to FIG. 30 above).

An inductor LI, a capacitor C1, main switching transistor QI arecontrolled by the operation of two switches SWA and SWB operating athigh speed of the order >I 0 KHz<I 00 MHz in two phases called Phase Aand Phase B. There is also a period where neither switch is activatedcalled Phase C. The numerical example here has F=2 MHz, a diode throughcurrent of 10 A t(Phase A)=450 nS:t(Phase B)=SO nS ratio i.e.forced-beta of 9.

QI is either a standard high-voltage BJT device operating inreverse-beta mode, or an optimal transistor structure such as the B2device. SWA, SWB can be BJT, Mosfet or Jfet or other low-voltage typedevices.

The node VSS is a convenient internal circuit node relative to whichother voltages are to be measured and can be one terminal of anauto-generated power supply for internal control circuits, the otherbeing VDD.

We assume that the start-up process is complete, the D2-device isforward biased by the circuit current and Inductor current is flowing atall times in LI i.e. continuous conduction mode (discontinuous mode isalso possible) and this current steered by switches SWA or SWB ontoeither the CE1 terminal (i.e. VSS) or the Base terminal of thetransistor. The inductor path is in the only DC conduction path throughthe ‘diode’ so it is generally apparent that its current must equal, onaverage, the load circuit current.

During phase A where SWA is on, the inductor current in LI builds upwith a delta-I component at a rate set by V(C 1) voltage (which is oforder 70 mV)/L 1, and will exceed the load current by the end of Phase Atime. FIG. 38C has some illustrative waveforms.

In phase B, where SWA is switched off and SWB is switched on, theinductor current generally continues but voltage will instantaneouslyboost to a voltage set by the BASE of Q I

-   -   typically 0.75V (plus a little for the voltage drop of SWB). The        inductor current droops during this time at a rate set by        (VBE-V(C1))/L1, dropping below the load current value.        Transistor QI has high, current-controlled, stored charge        represented by Chase and slow turnoff speed such that these        discontinuous BASE current pulses are averaged into an effective        base current within the transistor. It can be seen that there is        no net charge given to Chase in a cycle where I A*450 nS=9 A*SO        nS.

The time-slicing of the inductor current between SWA and SWB paths intoCEI and BASE terminals respectively gives a precise ‘Forced-Beta’ ratioto QI of t(PhaseA):t(PhaseB)

This ratio is maintained over all load currents without need for controlsystem intervention and thereby keeps the transistor optimally driven.

It was noted that the inductor sees a ripple current exceeding thendropping below the load-current value during the switching cycle. Sincethe load current value doesn't change over the short switching cycle, itfollows that the L 1 ripple current must be balanced by an equal ripplecurrent to/from C 1. C1 is therefore critical and is scaled up inproportion to expected inductor ripple current (which itself isinversely proportional to the chosen L 1 inductor value), scaled up toreduce voltage ripple at the external terminals of the D2-device, andscaled down as the operating frequency increases.

Typical values for a 10 A ‘diode’ switching at 1 MHz are L1 =10 nH, C1=47 uF, Cbase=22 uF (using standard BJT). Chase is generally not neededwhen using a B2 type device typically operating at 100 A/cm2 currentdensity and a forced-beta of 9. It is important to note that C1 only hasto be rated at a few volts and can be a very small ceramic typecapacitor. Inductor L1 is of such small value that it can occurnaturally from ‘ parasitic’ bond-wire inductance or PCB trace and neednot use additional magnetic materials.

Effective Vf of the ‘Diode’:

The object of this embodiment is to make a diode with a much lowerforward voltage than a standard PN junction diode.

Ignoring the small losses from parasitic resistances, the effectiveforward voltage of the ‘diode’ are made up of two components. 1) theaverage voltage over C 1 plus 2) VCE(sat) of Q1.

The first component can be calculated from the voltage boost ratiobetween V(C 1) and VBE of Q1. The voltage boost ratio is (t(PhaseA)/t(Phase B))+1, because of what is essentially a boost converteroperating with L 1, SWA, SWB and the +1 resulting from the fact that V(C1) is additive with the inductor boost voltage for driving VBE. SinceVBE is typically 0.75V, and if the t(Phase A):t(Phase B) ratio [i.e.forced-beta] is chosen to be 9:1 then C1 will settle to 75 mV when theconverter reaches equilibrium i.e. no net change of inductor currentover a complete cycle. Adding the typical 50 mV VCE(sat) of Q1 gives atotal ‘diode’ Vf drop of 0.125V in this example.

Vf can be reduced further by running with higher forced-beta settings(requires higher Beta of the transistor) and lower VCE(sat) which comesfrom using larger transistor operating at lower current density.Thinned-wafer B2 devices designed for 650V breakdown exhibit Betas ofbetween 30 and 10.

Reverse Blocking Characteristic:

In reverse-bias, and with C 1 discharged, Q 1 acts as a high voltage‘off NPN transistor with its Base biased to its Emitter by the leakageresistance of the active circuitry, or extra ‘pull-down’ resistanceadded if required. The reverse breakdown voltage is fully thatdetermined by the design of Q1.

Self-Oscillating Version:

FIG. 37 D,E,F describe a working system which uses self-sustainingoscillation provided by the addition of coupled-inductor L2 and twolow-voltage discrete BJT devices Q2, Q3 to implement SWA and SWB. An LCoscillator is formed from the coupling of L 1 and L2 where a typical 9:1turns ratio can in principle produce a 0.75V Vbe to turn on Q2 base whenV(C 1) is only 75 mV. This circuit has positive feedback and oscillatesnaturally between Phase A and Phase B operation as voltage C 1 ripplesup and down at a rate set by L 1. L3 here is used to reduce the forwardvoltage of Q3 which is acting as SWB and could equally be replaced byother rectifier means such as Schottky diode, PNP with grounded baseeither of which method would no longer require L3. If VBE of Q2 issimilar to VBE of Q 1 then the PWM duty cycle is similar to the L2:L1turns ratio.

Integrated SWA, SWB and Power transistor for self-oscillation version.

FIG. 37C is a cross section of a device which incorporates Q1 as avertical high voltage power device with Q2 and Q3 as low-voltagelateral/vertical devices giving a monolithic semiconductor solution.

NMOS Vs. BJT for SWA, SWB

Using BJTs for SWA and SWB limits the ultimate forward voltage of theD2-device to around 200 mV because BJTs do not quite saturate down to 0Vas do MOS devices. Using NMOS devices, perhaps in another die to Q1, andusing more accurate pulse timing than can be achieved withself-oscillation can bring the total loss to around 100 mV at low cost.

Driving SWA and SWB from a more sophisticated control circuit than asimple oscillator has advantages for the ‘diode’ application, butparticularly for Transistor/Thyristor type applications and especiallyfor smart-power devices.

C2-Device

FIG. 38A shows the concept of a device called C2 with the C referring toCMOS processing which would be typically used although NMOS processcould be used for lower cost.

Although this could be built from two separate die—a CMOS controller ICmounted on top of or alongside an I2 switch, FIG. 38B,D have thepreferred embodiment where the 12 devices if formed underneath true CMOSstructures on a monolithic wafer. This means that all

types of circuits and IP blocks can be used to make a self-powered,smart-power transistor with fully built-in driver electronics e.g. ADC,DAC, Flash Memory, RAM, Microprocessor. An isolated control interfacecan be added using an off or on-die transformer making an easy to usesmart transistor.

SWA and SWB are NMOS devices which on a typical 0.18 u CMOS process havea specific on resistance of 0.5 m0hm*mm2. To handle 10 A, only I mm2 ofsilicon area is needed to give a negligible 5 millivolts of DC voltagedrop in SWA

A digital controller drives the gates of SWA and SWB running from a VDDsupply of typically 1.8 volts and switching at around 2 MHz. VDD isgenerated readily using Phase C period shown on the waveforms of FIG.38C as the short spike on the LI voltage trace. A VDD voltage regulatorfunction can be completed by control loop using on-chip components toadjust the time period or occurrence rate of the Phase C period.

Rdamp is the combination of real and parasitic resistance designed todampen oscillations as the PWM forced-beta ratio is changed.

Further Refinements to the Control Techniques:

First order BJT base current control is automatic—the base current isalways a fixed fraction of collector current over all possible collectorcurrent range—set precisely by the PWM ratio giving the Forced-beta.

It is possible to add a second-order loop which increases the PWM ratioin proportion to the measured collector current. This can account forthe roll-off in Beta with increasing collector current of sometransistor types. For high-speed low-loss turn off it has been foundbest to reduce the force-beta to a value below the sustaining value andonly shutting of base current when the VCE ICE2 voltage is seen to rise,rather than simply cutting off base current immediately.

Refinements Made Easier with CMOS Integration:

With multiple stripe construction and very fine granularity oftransistor formation a C2-device can contain vertical I2 transistorformations of different characteristics on the same die. Dramaticallydifferent characteristics result from changing the backside emitter frombeing opaque to being transparent to holes using multiple maskedimplants at the rear of the device. FIG. 38B shows a slow device besidea fast device.

NMOS devices SWB can be doubled-up to give independent Base_slow andBase_fast destinations for the base current. This technique will combinethe best of high-Beta, low VCE(sat) characteristic of the slow devicewith the low Eoff losses of the fast device by arranging for a two-stageturn-off where first the slow transistor is turned off, handoff ispassed

over by first turning on the fast transistor before this is turned offto exploit it very low Eoff characteristic.

In practice there would be larger segregation distances between the fastand slow devices—ideally the carrier diffusion length to reducecross-coupling. Also, the extremes of fully opaque and fully transparentmight not be used.

Start-Up Operation:

The self-oscillating ‘diode’ of FIG. 37D starts up within 10 uS of beingdriven in the forward direction once Q3 and QI PN junction voltages areexceeded (1.5V approx). Then C1 is quickly charged to around IV andoscillation begins strongly. This initially higher forward voltagepersists for such a short time relative to the 1000.times. longer 50 Hzhalf-cycle time that it can be ignored for mains rectificationapplication

C2-Device Start-Up and Standby:

The C2-device must operate at up to 100 KHz but an initial delay of evena few seconds from system power-up is acceptable. Using either thetransistor's inherent leakage current or deliberate leakage path asformed by high-voltage JFET structure can charge capacitor CI from thistiny current. With the extremely low standby power consumption inherentin CMOS, V(C 1) can ramp to 2.5V which will give an initial VDD of 1.8V(one diode drop—see FIG. 38A).

This represents a standby condition where the energy stored on C I isenough to begin which the device is ready to come into operational modeas soon as an input signal is received. Operational mode is as describedearlier with Phase A, B, C and self-powering technique. Dropping backinto standby mode means the device is ready to turn on again wheneverrequired without delay.

Advanced Bridge Rectifier with Value Added Features:

FIG. 39 shows ideas for combining active diodes and together to producefirst a basic low-loss bridge rectifier then an more advanced versionwith several added features

1) Overvoltage protection for downstream components. In the version ofFIG. 39B, an on-chip comparator on the C1VIOS die detects a mainsovervoltage and where the controlled transistor is a T2 device (asdescribed above), it can be tuned off, protecting any downstreamelectronics including electrolytic capacitors and power transistors fromovervoltage. These components can then be rated only for the typically450V excursions rather than rated at 650V device for infrequent mainsover voltages. Either cost savings, performance improvement and/orreliability increases can be expected from lower voltage, less stresseddevices. 2) Soft start

Normally an additional device such as NTC thermistor is used in mainsrectification applications to reduce inrush currents into theelectrolytic capacitors at the expense of wasted heat in the NTCthermistor after the inrush event. The active bridge T2 type devices caninstead give a pulse-width modulated gradual charging of theelectrolytic during power-up before switching to the normallow-forward-voltage mode.

3) Auxiliary Power Supply Output

A low voltage auxiliary power supply can be tapped from the internal VDDsupply and passed out of the bridge-rectifier module for use by externalcircuits such as Switch Mode Power Supply Controller or Power FactorController. These pins can be seen in FIG. 39D and FIG. 39E but are notbeing used in the example given.

FIG. 40 illustrates a metal assisted chemical etching process using amoving platform. In an embodiment, a Metal-Insulator-Semiconductor(M-I-S) Anodic etching and/or Metal assisted-chemical-etching using amoving platform 4000 is illustrated. For the understanding/explanationpurpose wherever a reference is made to a Metal, generally any conductorwould suffice as an alternative.

FIG. 40 shows a platform 4000 include cutting tools 4002, in the form ofarray of micro needles, connected to a nanometer accurate XYZpositioning table adapted to oscillate in X-Y Z axis directions. Theplatform 4000 has a bearings 4004 connected to a support 4006 allowingthe positioning table to freely oscillate. A substrate 4008, preferablea silicon wafer, is placed in a container is positioned under the arrayof micro-needles 4002. The container is filled with an acidic solution4010, preferably Hydrofluoric (HF) acid. A lamp 4012 may be provided toilluminate backside of the substrate 4008.

In an embodiment, a method of etching a substrate, wherein the substrateis a silicon substrate or a substrate having a silicon surface, isdisclosed. The method includes placing the substrate in a container,wherein the substrate is a N-type substrate; providing a volume of anacid solution in the container, wherein the acid solution serves as aninsulator; and drilling, using one or more needles supplied with avoltage 4014, one or more holes on the surface of the substrate tolocally invert the N-type substrate to a P-type substrate, wherein thevoltage applied on the surface of the substrate anodically etches thesurface of the substrate to create the one or more holes by surfaceinversion.

In an embodiment, as shown in FIG. 40, an anodic etching is achieved byusing needles, for example an array of micro-needles. Making the needlesnegative makes them act like metal gates in a MOS structure. In anexemplary implementation, a wafer to be etched is N-type and a liquidserves as an insulator. With a voltage of say 10 volts, when the needlegets within about

500 nm of the silicon surface the silicon will locally invert fromN-type to P-type and holes will appear at the surface of the silicon.Because of the applied circuit voltage the surface of the silicon willbe anodically etched—similar to the way microporus silicon can be formedbut in this case holes are created by surface inversion. The microneedlearray is positioned on a nanometer accurate XYZ positioning table. Asmall oscillating X/Y movement of perhaps +/−1OO nm can serve asmechanical agitation of the etching solution although it will make theetched features this much larger. Z oscillating motion can be used toform a piston-type pumping effect to further improve circulation ofetchant and effluent to maintain high etch rate.

In another embodiment, a high frequency possibly bipolar electricalpulse rather than DC may reduce conduction losses in the Hydrofluoric(HF) acid and only allow for short pulses where the inversion layer isbeing etched compared to long times when it is recovering.

In an embodiment, the etching rate is set by the current which in turndepends on the surface area being etched.

It would be appreciated that, this system could also be applied to aMetal-assisted chemical version which mechanically would work similar tothat set out for M-I-S etch but now the micro-needles would be platedwith noble metal and the etching solution would be HF

+H202 and no electrical supply would be required.

Cost Estimates/Cost of Ownership

Standard Bridge rectifiers are very inexpensive devices, costing around$0.50 for a 10 A 600V plastic packaged bridge rectifier. This splits70%:30% between the cost of the diodes and the assembly/packagingmaterials.

However, the 15 watts minimum it will lose as heat costs incurs andextra $1 to deploy the necessary heatsinking means. Additionally, thecost of ownership is around $15 per year from electricity cost to fuelthe 15 W of heat it generates [based on mean world electricity price of$0.19 per KwHr]

D2-device bridge rectifier. 10 A bridge-rectifier made up according toFIG. 3E has the following costs.

CMOS device. Assume $800 per 8″ 24-mask wafer and kerf losses @1.5mm.times.1.5 mm=$0.057.times.4=$0.23

B2 device. Assume $200 per 8″ two-mask process wafer @3.3 mm.times.3.3mm=$0.064.times.4=$0.255

Packaging+50% —package will have profiled shape to aid heat dissipation.

Testing/Yield losses+20%>

Total=.about.$0.87

The price is competitive and is actually cheaper than the standardbridge rectifier if the price of its heatsink is included. Initial costcomparisons however are mute since using a D2-device bridges will saveits owner $12.50 per year per bridge at 10 A of rectified current.Payback time is in the order of 2 weeks. It also adds more than 1% tothe efficiency figure of a 120 Vac mains operated device which is animportant marketing metric for appliance manufacturers

It will be appreciated that the invention can be described in thefollowing clauses:

1. A power switching semiconductor device of bipolar construction and ofPNP or NPN structure where minority carriers are injected from a baseterminal which is normally kept inoperative and out-of-circuit by adepletion region induced by adjacent diffusion of opposite polarity andhigher doping or by a junction transistor formed where there isencroachment of the adjacent diffusion or a pre-deposited junction.2. A power switching semiconductor device according to clause 1 wherehigh reverse bias voltages are supported by a lightly doped drift regionwhich comprises the Base of the transistor.3. A power switching semiconductor device according to any precedingclause where conduction occurs in two quadrants supporting operation asan AC switch.4. A power switching semiconductor device according to any precedingclause which has two base connections, one on each side of the wafer, tosupport efficient AC switching gam.5. A power switching semiconductor device according to any precedingclause which has one base connection on one side of the wafer which isdriven by a single DC base supply.6. A power switching semiconductor device according to any precedingclause with microprocessor controlled buck regulation of the basevoltage/current drive and analogue to digital feedback information intothe microprocessor's algorithms.7. A power switching semiconductor device of PNP or NPN constructionwhere minority carriers are injected from a base terminal which isnormally kept inoperative and out of-circuit by a depletion regioninduced by adjacent diffusion of opposite polarity and higher dopingwhere minority carriers are injected on the upper part of thesemiconductor using a dedicated emitter terminal and where twocollectors displaced laterally are formed on the lower side of thesemiconductor to form the switching terminals.8. A power switching semiconductor device according to any precedingclause where the energy used to power the base of the device is derivedfrom the conduction voltage drop of the device.9. A power switching semiconductor device according to any precedingclause where emitter/collector regions are etched then subsequentlydiffused with dopant.10. A power switching semiconductor device according to any precedingclause where Quasi-PNP (for overall NPN device) or NPN (for overall PNPconstruction)/Quasi JFET electrode structure is present.11. A power switching semiconductor device according to any precedingclause where conduction is synchronised to mains voltage cycle waveformfor zero-crossing switching.12. A power switching semiconductor device according to any precedingclause used to synchronously rectify mains AC power to reduce powerlosses compared to the Vf of a standard semi conductor diode.13. A power switching semiconductor device according to any precedingclause where Polysilicon trench fill is used to form emitter/collectorsof high doping and/or thin inter-facial oxide feature.14. A power switching semiconductor device according to any precedingclause where collector/emitters suffer reduced minority carrierinjection through means of Silicon Germanium or other electric fieldgrading technique.15. A power switching semiconductor device according to any precedingclause forming 3 d or stacked devices to give higher power abilityand/or higher sensitivity and lower conduction losses.16. A power switching semiconductor device according to any precedingclause where the stacked structure is used to increase the surface areaof the finished article to obviate the need or a dedicated heat-sink.17. A power switching semiconductor device according to any precedingclause which incorporates a charge-control model of bipolar diffusioncurrent transport within its algorithms.18. A power switching semiconductor device according to any precedingclause which is self-powered from the load using an auxiliary transistortap circuit switching on around the zero-crossing times.19. A power switching semiconductor device according to any precedingclause where the finished article is calibrated and coefficients storedin non-volatile memory mounted with the transistor.20. A power switching semiconductor device according to any precedingclause using a recessed BASE contact (auxiliary transistor emitter) sothat the a CE electrode is prominent

allowing 3D device stacking with interspaced conductor sheets for heatextraction and electrical conduction.

21. A power switching semiconductor device according to any precedingclause using a transformer coupling arrangement of LF power waveform andRF data waveform allowing for data networking and isolated power betweenan number of intelligent nodes.

22. A power switching semiconductor device according to any precedingclause where the required control system can be powered using theinbuilt boost converter acting on the energy available from the throughcurrent of the main semiconductor switch.

23. A power switching semiconductor module according to any precedingclause acting in the overall capacity as a two terminal fuse for acircuit to be protected.

24. A power switching semiconductor with a current-mode base controlwhere resistive DAC or DACs are used to control current of the overallor individual bases of the power semiconductor according to a controlprogram reactive to the measured operating conditions of the device.25. An array of power switching semiconductors according to anyproceeding clause combined with a multichannel control circuit to effecta matrix converting ac to ac power converter.26. A power switching semiconductor according to any preceding clausewhere a buffer layer is inserted of opposite doping under the CE2 orCollector terminal and has doping around the standard concentrationranges typical for punch-through or filed-stop control layers used inIGBTs or Diodes.27. A power switch driver comprising a coupled dual PWM system where thelatter is a high-frequency buck-mode converter using inductor and theformer is a standard PWM channel and where the cross modulation of thetwo PWMs across a base drive inductance affords a continuous dynamiccurrent control of a power switch with a high speed on and off ability.28. A coupled dual PWM driver where PWM2 frequency is such that the offtime is of similar order or not much longer than the minority carrierlifetime ensuring that conductivity of the switched transistor remainsapproximately constant in the PWM2 period.29. Multiple, independently programmable, copies of the dual PWM drivereach driving a separate finger or power device die to provide theability to control current and timing independently on each region of adie or multiple die.30. A coupled multi-channel driver where multiple phase-offset PWM2 typedrivers each driving an inductor with a common point on the baseterminal of a transistor and a single PWM1 channel to drive standarddevices such as silicon-carbide NPN transistor.31. A coupled multi-channel driver according to above where multiphaseoffsets create a higher effective PWM2 frequency to match the reducedminority-carrier lifetime of high speed transistors such assilicon-carbide.32. An intelligent power-transistor driver with multi-channel outputs asdescribed in previous clauses, which applies during operationpre-programmed co-efficient which had been determined aftermanufacturing to make a device with the overall uniform device voltage,current and temporal characteristics of a single device.33. An intelligent power-transistor driver with multi-channel outputs asdescribed in previous clauses with an adaptive circuit to respond to anexternal PWM and by applying known delay coefficients and calibrationvalues known in advance to create the same PWM on/off ratios at theoutput transistor albeit with a slight overall delay.34. An intelligent low leakage relay circuit of construction accordingto any preceding clause where a second switch shunts any leakage currentaround the load during switch off and optionally a third transistordisconnects to achieve Pico-ampere level leakage currents into the load.35. An AC switchable power transistor and driver combination accordingto any preceding clause where a virtual diode action is formed by thedriver being commanded or detecting the need for reverse conduction andturning on the transistor accordingly.36. A minority-carrier switching transistor whose base terminal andinternal structure current rating equal to that of the collector oremitter current rating so that it yields a reverse free-wheeling diodefunction of equal current rating to the forward rating and when the baseis actively clamped to ground using typically an NMOS FET from thedriver or when the base current is produced by the driver to effectreverse bias switch-on transistor action and to further reduce thevoltage of the C to E path below the normal Vdiode drop.37. A High-side DC switch or multiple high side DC switches comprising atransistor according to any preceding clause and a single ormultichannel driver according to any preceding clause

It will also be appreciated that still further aspects of the inventioncan be described in the following further clauses:

According to an aspect, a bi-directional bipolar junction transistor(BJT) structure is provided, including comprising: a base region of afirst conductivity type, wherein said base region constitutes a driftregion of said structure; first and second collector/emitter (CE)regions, each of a second conductivity type adjacent opposite ends ofsaid base region; wherein said base region is lightly doped relative tosaid collector/emitter regions; the structure further comprising: a baseconnection to said base region, wherein said base connection is withinor adjacent to said first collector/emitter region.

According to an aspect, said first and second collector/emitter regionsand said base region define a bi directional BJT, and wherein aconnection to said base region of said bi-directional BJT via a secondtransistor having a first input/output (I/O) terminal connected to saidbase connection, and a control connection coupled to said first CEregion.

According to an aspect, said second transistor is ajunction gatefield-effect transistor (JFET), wherein said base connection and baseregion are source/drain connections of said JFET, wherein said controlconnection is a gate terminal of said JFET, wherein said base connectionis adjacent said first CE region, and wherein a channel region of saidJFET is between said base connection and said base region.

According to an aspect, said second transistor is a BJT, wherein saidfirst I/O terminal has said first conductivity type, wherein said baseconnection is within said first CE region, and wherein said controlconnection of said BJT is a base is formed by a portion of said first CEregion.

According to an aspect, said base region is wider in a direction betweensaid ends of said base region than each of said collector/emitterregions, and wherein a current carrying capability of a connection pathbetween said base connection and said second CE region is less than acurrent carrying capability of a connection path between said first andsecond CE regions.

According to an aspect, a forward conduction path from said base regionto said second CE region is driven by a voltage on said base regionrelative to said second CE region, and wherein, when said forwardconduction path is present, a forward conduction path between said baseconnection and said base region includes a depleted portion of said baseregion.

According to an aspect, when no voltage is applied to any terminals, thestructure is in an off-state so as to form depletion regions betweensaid first CE region and base region and between said second CE regionand base region.

According to an aspect, when a positive voltage is applied to saidsecond CE region and no voltage is applied to the first CE region andthe base connection, the structure is in an off-state so as to form adepletion region between said second CE region and base region.

According to an aspect, when a negative voltage is applied to saidsecond CE region and no voltage is applied to the first CE region andthe base connection, the structure is in an off-state so as to form adepletion region between said first CE region and base region.

According to an aspect, when a first positive voltage is applied to saidsecond CE region, a second positive voltage being applied to the baseconnection and no voltage is applied to the first CE region, thestructure is in an on-state in which majority carriers from the first CEregion flow through the base

region towards the second CE region, and minority carriers from the baseconnection are injected into the base region, the minority carriersbeing recombined with the majority carriers in a region adjacent thefirst CE region.

According to an aspect, when a negative voltage is applied to saidsecond CE region, a positive voltage being applied to the baseconnection and no voltage is applied to the first CE region, thestructure is in an on-state in which majority carriers from the secondCE region flow through the base region towards the first CE region, andminority carriers from the base connection are injected into the baseregion flowing towards the second CE region, the minority carriers beingrecombined with the majority carriers in a region adjacent the second CEregion.

According to an aspect, the first conductivity type comprises a p-typedoping polarity and the second conductivity type comprises an n-typedoping polarity.

According to an aspect, said structure is non-latching and switches offa connection between said first and second CE regions on removal of avoltage from said base connection.

According to an aspect, said base connection is recessed into a surfaceof said structure.

According to an aspect, said base connection is an ohmic connectioncomprising a region of said first conductivity type, and wherein saidbase region is of ohmic type.

According to an aspect, said ohmic base region is configured to drivetransistor comprising the first CE region, base region and second CEregion into a saturation region during current conduction.

According to an aspect, said device is a vertical device. According toan aspect, said structure is a lateral structure.

According to an aspect, the second CE region comprises two separationportions laterally disposed to one another and wherein each separateportion forms a switching terminal.

According to an aspect, a bipolar junction transistor (BJT) structure,comprising: a base region of a first conductivity type, wherein saidbase region constitutes a drift region of said structure; first andsecond collector/emitter (CE) regions, each of a second conductivitytype adjacent opposite ends of said base region; wherein said baseregion is lightly doped relative to said collector/emitter regions; thestructure further comprising: a base connection to said base region,wherein said base connection is within or adjacent to said firstcollector/emitter region and a buried layer of the second conductivitytype disposed between the second CE region and the base region.

According to an aspect, the structure is configured to operate in a DCapplication.

According to an aspect, a bipolar junction transistor (BJT) structure,comprising: a base region of a first conductivity type, wherein saidbase region constitutes a drift region of said structure, the driftregion being a reverse voltage sustaining region; a collector region ofa second conductivity type; an emitter of a second conductivity type,the collector and emitter being adjacent opposite ends of said baseregion; wherein said base region is lightly doped relative to saidcollector and emitter regions; the structure further comprising: a baseconnection region of the first conductivity type formed adjacent to saidemitter region and a field stop layer of the first conductivity typeformed between the emitter region and the base region, the baseconnection being within the field stop layer.

According to an aspect, the doping concentration of the field stop layeris less than that of the base connection.

According to an aspect, the thickness of the field stop layer is morethan that of the base connection. According to an aspect, that a diodeis formed between the collector and base region.

According to an aspect, the diode is configured to operate as a reverseconducting diode when driven by a driver circuit.

According to an aspect, the driver circuit comprising a first PWMcontroller and a second PWM controller, the first and second PWMcontrollers being coupled to one another, wherein the first PWMcontroller is capable of controlling a low voltage transistor and thesecond PWM controller is a high frequency converter.

According to an aspect, the second PWM controller is a buck converterusing an inductor.

According to an aspect, the cross modulation of the first and second PWMcontrollers across a base drive inductance allows a continuous dynamiccurrent control of the BJT structure with high speed on and offcapability.

According to an aspect, the frequency of the second PWM controller issuch that the off time is of substantially similar order compared to theminority carrier (e.g. electron) lifetime ensuring that conductivity ofthe BJT structure remains substantially constant during the periodcontrolled by the second PWM controller.

According to an aspect, the second PWM controller is a phase offsetcontroller.

According to an aspect, a third PWM controller which is a phase offsetcontroller, wherein the second and third phase offset controllers eachdrive an inductor with a common point on a base connection terminal of atransistor.

According to an aspect, the first PWM controller is coupled to the baseconnection terminal of the transistor so as to drive the transistor.

According to an aspect, the multiphase offsets of the second and thirdPWM controllers create a relatively high effective frequency to match areduced minority-carrier lifetime of high speed transistors.

According to an aspect, the combination of the driver circuit and theBJT structure is configured to provide a reverse conducting diode.

According to an aspect, the base region is of p conductivity type, thedriver is configured to detect a negative current of the base so thatholes can be pulled out from the junction between the first CE regionand base to clamp a negative excursion.

According to an aspect, the current rating of the base regionsubstantially equal to that of the collector or emitter so as to yield areverse free-wheeling diode of substantially equal current rating to theforward rating.

According to an aspect, the base region is configured to be clamped toground using a MOSFET from the driver.

According to an aspect, the base current is produced by the drivercircuit to result in a reverse bias switch-on transistor action so as toreduce the voltage drop in a current flow path below a normal voltagedrop of a diode.

According to an aspect, a computer program product comprising a computerreadable medium in which a computer program is stored, the computerprogram comprising computer readable code which, when run by acontroller of a driver circuit, causes the driver circuit to operate asthe driver circuit, wherein the computer program is stored on thecomputer readable medium.

According to an aspect, when the controller is configured to detect avoltage of the BJT structure in an off-state, the computer programproduct is configured to turn on the BJT structure to emulate a reverseconducting diode action.

According to an aspect, a driver circuit, in particular for a BJTstructure driver, comprising: i) a voltage sensing resistor and ii) acurrent sensing resistor, each coupled to one of said CE regions, and amicrocontroller coupled to receive a voltage sensing signal and acurrent sensing signal from said resistors and to provide a PWM outputfor controlling a base current into said base connection via aninductance.

According to an aspect, a circuit breaker comprising a first, powersemiconductor switching device and a driver circuit, wherein saidcircuit breaker has two power switching terminals, and further comprisesa power supply, and a controller for said power semiconductor switchingdevice powered by said power supply, wherein said power supply iscoupled in series with said first power semiconductor switching devicebetween said power switching terminals to derive a power supply fromsaid terminals whilst said power semiconductor switching device is on,and wherein said power supply comprises a second switching devicecoupled in series with said first power semiconductor switching device,between said power switching terminals, such that the circuit breaker isoperable without a separate power supply. According to an aspect, saidfirst device is a high voltage device, said second device is a lowvoltage device comprising part of a de-to-de power convertor inputstage.

According to an aspect, said power supply is a switched mode powerconvertor comprising a plurality of low voltage switching devices tocharge and discharge an energy storage component such that a powersupply for said controller is provided irrespective of the direction ofcurrent flow between said power switching terminals, the circuit breakerfurther comprising a sensor to sense said direction of said current flowfor controlling said plurality of switching devices.

According to an aspect, a circuit breaker further comprises a reservoircapacitor charged by leakage current through said power switchingdevice, to provide a power supply for sensing said direction of currentflow at start-up of said power supply.

According to an aspect, a circuit breaker operably connected to the BJTstructure, the circuit breaker comprising: an input capacitor connectedto a CE region; an inductor coupled to the input capacitor;

first and second switching devices coupled to the inductor; a secondcapacitor coupled to the second switching device; and a pulse widthmodulation (PWM) controller configured to control the first and secondswitching devices.

According to an aspect, when a positive voltage is applied to the CEterminal, the first switching device is configured to charge theinductor, and the second switching device is configured to charge thesecond capacitor.

According to an aspect, the charging of the inductor is controlled bycontrolling the duty cycles of the PWM controller.

According to an aspect, when a negative voltage is applied to the firstCE terminal, the second switching device and the second capacitor aredisconnected from the circuit breaker.

According to an aspect, a third switching device and a third capacitorwhich are coupled to the first switching device, the inductor and thefirst capacitor.

According to an aspect, the third switch is configured to charge thethird capacitor.

According to an aspect, a bootstrap circuit operatively connected to theBJT structure and operatively connected to the circuit breaker, thebootstrap circuit comprising a first diode coupled with the secondcapacitor of the circuit breaker and a second diode coupled with thethird capacitor of the circuit breaker, wherein the bootstrap circuit isconfigured to store positive or negative leakage current in the firstand/or third capacitors through the first and second diodes so as toturn on the bi-direction BJT structure.

According to an aspect, a bleed resistor to provide sufficient currentto turn on the BJT structure if there is inherent leakage currentpresent in the BJT structure.

According to an aspect, an auxiliary tap circuit switching on around thezero-crossing times so as to power the BJT structure.

According to an aspect, a driver circuit operatively connected to aplurality of BJT structures, wherein each BJT structure is disposed sideby side on a chip and wherein the driver circuit comprises a pluralityof independent PWM drivers each independently driving the baseconnection of each BJT structure through an inductor.

According to an aspect, each PWM driver is configured to control currentto the base connection and switching time of the BJT structureindependently.

According to an aspect, each PWM driver is configured to control thecurrent during an on-state of the BJT structure using a discontinuouscurrent inductor drive.

According to an aspect, the discontinuous current mode occurs when anoff-time from the PWM driver is sufficiently long so that the inductorcurrent decreases to zero.

According to an aspect, a driver circuit operatively connected to a BJTstructures, comprising a resistive digital to analogue controller (DAC)for controlling the current of the base of the BJT structure.

According to an aspect, the DAC is configured to control the basecurrent ofthe BJT structure according to a control program which isreactive to measured operating conditions of the BJT structure.

According to an aspect, a matrix converter comprising an array of BJTstructures, the matrix converter further comprising a control circuitcomprising a plurality of channels which are configured to control theswitching of the array of BJT structures.

According to an aspect, a relay circuit for a low leakage currentapplication, the relay circuit comprising the BJT structure, the relaycircuit further comprising a load resistor and a switching devicearranged parallel to the load resistor, wherein the switching device isconfigured to bypass any leakage current from the BJT structure aroundthe load resistor during switching off operation.

6 According to an aspect, a further switching device coupled with theload resistor, the further Switching device being configured to obtainPico-ampere level leakage current into the load resistor.

According to an aspect, a driver chip operatively connected to a BJTstructures and comprising the driver circuit, wherein the driver chip isconfigured to apply pre-programmed coefficients determined aftermanufacturing the components of the driver chip.

According to an aspect, a driver chip, wherein the first PWM controlleris configured to vary phases for different regions of the BJT structurebased on calibration parameters of the driver chip so as to allow alarge die including the BJT structure to turn on and/or off tocompensate for the difference in for example carrier lifetime and/ordoping levels.

According to an aspect, a driver chip, wherein the driver chip ismounted directly on top of a wafer comprising the BJT structure.

According to an aspect, a method of manufacturing a bipolar junctiontransistor (BJT) structure, the method comprising: forming a base regionof a first conductivity type, wherein said base region constitutes adrift region of said structure; forming first and secondcollector/emitter (CE) regions, each of a second conductivity typeadjacent opposite ends of said base region, wherein said base region islightly doped relative to said collector/emitter regions; and forming abase collection to said base region, wherein said base connection iswithin or adjacent to said first collector/emitter region.

According to an aspect, a method according to clause 69, furthercomprising: etching the first collector/emitter region; and forming adopant diffusion in the etched region.

According to an aspect, filling polysilicon in a trench to form thefirst collector/emitter region and/or to form a thin interfacial oxideregion.

According to an aspect, applying an anisotropic wet chemical etching ofthe first collector/emitter region with artwork aligned at either zerodegrees or 45 degrees to form a simultaneous undercut of an oxide and aself-terminating V-groove etch of contact holes.

According to an aspect, applying the anisotropic wet etching to form abevel etch to control the edges of the BJT structure.

According to an aspect, applying an electric field grading technique toreduce minority carrier injection from the collector/emitter regions.

According to an aspect, forming a three dimensional or stacked structureso as to give higher power ability and/or higher sensitivity and lowerconduction losses.

According to an aspect, forming a recessed BASE contact so that theelectrodes on collector/emitter regions can form the three dimensionalor stacked structure.

According to an aspect, an active rectifier, comprising: a power bipolarjunction transistor (BJT), having a first and second input/output (I/O)connections and a base connection; first and second rectifier terminals,wherein said first I/O connection of said BJT is coupled to said firstrectifier termlinal, wherein said second I/O connection of said BJT iscoupled to said second rectifier terminal; a driver oscillator toprovide a two phase drive waveform having a first (on) portion and asecond (oft) portion; at least one controllable switch controlled bysaid driver oscillator and coupled between said second rectifierterminal, said base connection of said BJT and said second I/Oconnection of said BJT, to selectively route current from said secondrectifier terminal between said second I/O connection of said BJT andsaid base connection of said BJT; wherein said driver oscillatorcontrols said controllable switch to route said current from said secondrectifier terminal between said base and second I/O connections of saidBJT in proportion of a ratio of durations of said first and secondportions of said drive waveform.

According to an aspect, said second I/O connection of said BJT iscoupled to said second rectifier terminal via a filter, and wherein saidfilter comprises a capacitor such that a connection between said secondI/O connection of said BJT and said second rectifier terminal is viasaid capacitor.

According to an aspect, an inductance between said second rectifierterminal and said base connection said BJT to store current for saidbase connection whilst said controllable switch is routing current fromsaid second rectifier terminal away from said base connection of saidBJT.

According to an aspect, said controllable switch comprises a firstcontrollable switch coupled between said second rectifier terminal andsaid second I/O connection of said BJT and a second controllable switchcoupled between said second rectifier terminal and said base connectionof said BJT and; and wherein said two phase drive waveform comprisesfirst and second waveforms, said first waveform having an on portioncorresponding to said first portion of said two phase drive waveform,said

second waveform having an off portion corresponding to said secondportion of said two phase drive waveform, wherein said first waveformcontrols said first controllable switch and said second waveformcontrols said second controllable switch.

According to an aspect, said controllable switch comprises a firstcontrollable switch coupled between said second rectifier terminal andsaid second I/O connection of said BJT and a second controllable switchcoupled between said second rectifier terminal and said base connectionof said BJT and; and wherein said two phase drive waveform comprisesfirst and second waveforms, said first waveform having an off portioncorresponding to said second portion of said two phase drive waveform,said second waveform having an on portion corresponding to said firstportion of said two phase drive waveform, wherein said first waveformcontrols said first controllable switch and said second waveformcontrols said second controllable switch.

According to an aspect, a boost converter to boost a voltage drop acrossone or more circuit elements coupled between said rectifier terminals toprovide a power supply for said drive oscillator.

According to an aspect, said boost converter is coupled across one ormore circuit elements coupled in an emitter circuit of said BJT

According to an aspect, an inductance between said second rectifierterminal and said base connection said BJT to store current for saidbase connection whilst said controllable switch is routing current fromsaid second rectifier terminal away from said base connection of saidBJT; and wherein said boost converter comprises said inductance, toboost said voltage drop, and said driver oscillator such that saiddriver oscillator, and inductance together with said at least onecontrollable switch from a boost converter to power said driveroscillator.

According to an aspect, an active rectifier configured to use leakagecurrent through said BJT, or a high voltage current source device, or aresistor, to provide power to bootstrap said driver oscillator of saidbooster converter.

According to an aspect, said portions of said first and second waveformsare non-overlapping such that there is a dead time between said onportions; the active rectifier further comprising a power harvestingdevice or Schottky diode coupled to a connection between said secondrectifier terminal and said second I/O terminal of said BJT to harvestpower from said voltage drop during said dead time.

According to an aspect, said first I/O connection of said BJT is acollector connection and said second I/O connection of said BJT is anemitter connection.

According to an aspect, said ratio of durations of said first portion tosaid second portion of said two phase drive waveform is less than 1:1.

According to an aspect, a bi-directional bipolar junction transistor(BJT) structure, comprising: a base region of a first conductivity type,wherein said base region constitutes a drift region of said structure;first and second collector/emitter (CE) regions, each of a secondconductivity type adjacent opposite ends of said base region; whereinsaid base region is lightly doped relative to said collector/emitterregions; the structure further comprising: a base connection to saidbase region, wherein said base connection is within or adjacent to saidfirst collector/emitter region.

According to an aspect, said first and second collector/emitter regionsand said base region define a bi-directional BJT, and wherein aconnection to said base region of said bi directional BJT via a secondtransistor having a first input/output (I/O) terminal connected to saidbase connection, and a control connection coupled to said first CEregion.

According to an aspect, said second transistor is a junction gatefield-effect transistor (JFET), wherein said base connection and baseregion are source/drain connections of said JFET, wherein said controlconnection is a gate terminal of said JFET, wherein said base connectionis adjacent said first CE region, and wherein a channel region of saidJFET is between said base connection and said base region.

According to an aspect, said second transistor is a BJT, wherein saidfirst I/O terminal has said first conductivity type, wherein said baseconnection is within said first CE region, and wherein said controlconnection of said BJT is a base is formed by a portion of said first CEregion.

According to an aspect, said base region is wider in a direction betweensaid ends of said base region than each of said collector/emitterregions, and wherein a current carrying capability of a connection pathbetween said base connection and said second CE region is less than acurrent carrying capability of a connection path between said first andsecond CE regions.

According to an aspect, a forward conduction path from said base regionto said second CE region is driven by a voltage on said base regionrelative to said second CE region, and wherein, when said forwardconduction path is present, a forward conduction path between said baseconnection and said base region includes a depleted portion of said baseregion.

According to an aspect, when no voltage is applied to any terminals, thestructure is in an off state so as to form depletion regions betweensaid first CE region and base region and between said second CE regionand base region.

According to an aspect, when a positive voltage is applied to saidsecond CE region and no voltage is applied to the first CE region andthe base connection, the structure is in an off-state so as to form adepletion region between said second CE region and base region.

According to an aspect, when a negative voltage is applied to saidsecond CE region and no voltage is applied to the first CE region andthe base connection, the structure is in an off-state so as to form adepletion region between said first CE region and base region.

According to an aspect, when a first positive voltage is applied to saidsecond CE region, a second positive voltage being applied to the baseconnection and no voltage is applied to the first CE region, thestructure is in an on-state in which majority carriers from the first CEregion flow through the base region towards the second CE region, andminority carriers from the base connection are injected into the baseregion, the minority carriers being recombined with the majoritycarriers in a region adjacent the first CE region.

According to an aspect, when a negative voltage is applied to saidsecond CE region, a positive voltage being applied to the baseconnection and no voltage is applied to the first CE region,

the structure is in an on-state in which majority carriers from thesecond CE region flow through the base region towards the first CEregion, and minority carriers from the base connection are injected intothe base region flowing towards the second CE region, the minoritycarriers being recombined with the majority carriers in a regionadjacent the second CE region.

According to an aspect, the first conductivity type comprises a p-typedoping polarity and the second conductivity type comprises an n-typedoping polarity; and/or wherein said structure is non-latching andswitches off a connection between said first and second CE regions onremoval of a voltage from said base connection; and/or wherein said baseconnection is recessed into a surface of said structure; and/or whereinsaid base connection is an ohmic connection comprising a region of saidfirst conductivity type, and wherein said base region is of ohmic type,wherein optionally said ohmic base region is configured to drivetransistor comprising the first CE region, base region and second CEregion into a saturation region during current conduction.

According to an aspect, said device is a vertical device. According toan aspect, said structure is a lateral structure.

According to an aspect, the second CE region comprises two separationportions laterally disposed to one another and wherein each separateportion forms a switching terminal. According to an aspect, a bipolarjunction transistor (BJT) structure, comprising: a base region of afirst conductivity type, wherein said base region constitutes a driftregion of said structure; first and second collector/emitter (CE)regions, each of a second conductivity type adjacent opposite ends ofsaid base region; wherein said base region is lightly doped relative tosaid collector/emitter regions; the structure further comprising: a baseconnection to said base region, wherein said base connection is withinor adjacent to said first collector/emitter region and a buried layer ofthe second conductivity type disposed between the second CE region andthe base region.

According to an aspect, the structure is configured to operate in a DCapplication.

According to an aspect, a bipolar junction transistor (BJT) structure,comprising: a base region of a first conductivity type, wherein saidbase region constitutes a drift region of said structure, the driftregion being a reverse voltage sustaining region; a collector region ofa second conductivity type; an emitter of a second conductivity type,the collector and emitter being adjacent opposite ends of said baseregion; wherein said base region is lightly doped relative to saidcollector and emitter regions; the structure further comprising: a baseconnection region of the first conductivity type formed adjacent to saidemitter region and a field stop layer of the first conductivity typeformed between the emitter region and the base region, the baseconnection being within the field stop layer.

According to an aspect, the doping concentration of the field stop layeris less than that of the base connection; and/or wherein the thicknessof the field stop layer is more than that of the base connection.

According to an aspect, a BJT structure, being configured such that adiode is formed between the collector and base region, wherein the diodeis optionally configured to operate as a reverse conducting diode whendriven by a driver circuit.

Although the invention has been described in terms of preferredembodiments as set forth above, it should be understood that theseembodiments are illustrative only and that the claims are not limited tothose embodiments. Those skilled in the art will be able to makemodifications and alternatives in view of the disclosure which arecontemplated as falling within the scope of the appended claims. Eachfeature disclosed or illustrated in the present specification may beincorporated in the invention, whether alone or in any appropriatecombination with any other feature disclosed or illustrated herein.

The invention claimed is:
 1. A method of driving a bidirectionalsemiconductor power transistor device, the method comprising: drivingthe semiconductor power transistor device on a high side of thetransistor that is at a higher potential than an opposing low side ofthe transistor, the high side of the transistor comprising a P+electrode and a N+ electrode; selectively steering current into eitherthe P+ electrode of the semiconductor power transistor device or intothe N+ electrode of the semiconductor power transistor device on thehigh side of the semiconductor power transistor device.
 2. The method ofclaim 1, wherein driving the semiconductor power transistor devicecomprises driving a gate of the semiconductor power transistor device.3. The method of claim 1, further comprising using a controller todetermine, through a voltage comparator, the side of the transistor thatis at the lowest potential and driving the semiconductor powertransistor device accordingly.
 4. The method of claim 3, wherein thecontroller drives the semiconductor power transistor device byoutputting a PWM signal to drive switches.
 5. The method of claim 4,wherein the switches comprise low on-resistance mosfets.
 6. The methodof claim 1, wherein the bidirectional semiconductor power transistordevice comprises a double sided bidirectional semiconductor powertransistor device.
 7. The method of claim 1, wherein the bidirectionalsemiconductor power transistor device comprises a bidirectionalBipolar-Mode JFET (BMJFET).
 8. The method of claim 7, wherein thebidirectional Bipolar-Mode JFET (BMJFET) comprises a double sidedbidirectional Bipolar-Mode JFET (BMJFET).
 9. The method of claim 7,wherein the bidirectional Bipolar-Mode JFET (BMJFET) comprises anultra-lateral single sided bidirectional Bipolar-Mode JFET (BMJFET). 10.The method of claim 1, wherein the bidirectional semiconductor powertransistor device comprises a Silicon Carbide semiconductorconstruction.
 11. The method of claim 1, wherein the bidirectionalsemiconductor power transistor device comprises N-type wafer.
 12. Anelectronic component comprising: a bidirectional semiconductor powertransistor device; and a switching construction configured to drive thebidirectional semiconductor power transistor device in the mannerrecited in claim
 1. 13. The electronic component of claim 12, whereinthe electronic component comprises an auxiliary resonant commutated poleconverter auxiliary switch.